Full duplex backhaul radio with MIMO antenna array

ABSTRACT

A intelligent backhaul radio have an advanced antenna system for use in PTP or PMP topologies. The antenna system provides a significant diversity benefit. Antenna configurations are disclosed that provide for increased transmitter to receiver isolation, adaptive polarization and MIMO transmission equalization. Adaptive optimization of transmission parameters based upon side information provided in the form of metric feedback from a far end receiver utilizing the antenna system is also disclosed.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a Continuation application of U.S. application Ser.No. 14/632,624, filed on Feb. 26, 2015, which is a Continuationapplication of U.S. application Ser. No. 14/336,958, filed on Jul. 21,2014 (U.S. Pat. No. 9,001,809), which is a Continuation application ofU.S. application Ser. No. 13/898,429, filed on May 20, 2013 (U.S. Pat.No. 8,824,442), which is a Continuation of U.S. application Ser. No.13/536,927, filed on Jun. 28, 2012 (U.S. Pat. No. 8,467,363), which is aContinuation-in-Part application of U.S. application Ser. No.13/371,366, filed on Feb. 10, 2012 (U.S. Pat. No. 8,311,023), which is aContinuation application of U.S. application Ser. No. 13/212,036, filedon Aug. 17, 2011, (U.S. Pat. No. 8,238,318), and the disclosures ofwhich are hereby incorporated herein by reference in their entireties.

BACKGROUND

1. Field

The present disclosure relates generally to data networking and inparticular to a backhaul radio for connecting remote edge accessnetworks to core networks and an associated antenna system.

2. Related Art

Data networking traffic has grown at approximately 100% per year forover 20 years and continues to grow at this pace. Only transport overoptical fiber has shown the ability to keep pace with thisever-increasing data networking demand for core data networks. Whiledeployment of optical fiber to an edge of the core data network would beadvantageous from a network performance perspective, it is oftenimpractical to connect all high bandwidth data networking points withoptical fiber at all times. Instead, connections to remote edge accessnetworks from core networks are often achieved with wireless radio,wireless infrared, and/or copper wireline technologies.

Radio, especially in the form of cellular or wireless local area network(WLAN) technologies, is particularly advantageous for supportingmobility of data networking devices. However, cellular base stations orWLAN access points inevitably become very high data bandwidth demandpoints that require continuous connectivity to an optical fiber corenetwork.

When data aggregation points, such as cellular base station sites, WLANaccess points, or other local area network (LAN) gateways, cannot bedirectly connected to a core optical fiber network, then an alternativeconnection, using, for example, wireless radio or copper wirelinetechnologies, must be used. Such connections are commonly referred to as“backhaul.”

Many cellular base stations deployed to date have used copper wirelinebackhaul technologies such as T1, E1, DSL, etc. when optical fiber isnot available at a given site. However, the recent generations of HSPA+and LTE cellular base stations have backhaul requirements of 100 Mb/s ormore, especially when multiple sectors and/or multiple mobile networkoperators per cell site are considered. WLAN access points commonly havesimilar data backhaul requirements. These backhaul requirements cannotbe practically satisfied at ranges of 300 m or more by existing copperwireline technologies. Even if LAN technologies such as Ethernet overmultiple dedicated twisted pair wiring or hybrid fiber/coax technologiessuch as cable modems are considered, it is impractical to backhaul atsuch data rates at these ranges (or at least without adding intermediaterepeater equipment). Moreover, to the extent that such special wiring(i.e., CAT 5/6 or coax) is not presently available at a remote edgeaccess network location; a new high capacity optical fiber isadvantageously installed instead of a new copper connection.

Rather than incur the large initial expense and time delay associatedwith bringing optical fiber to every new location, it has been common tobackhaul cell sites, WLAN hotspots, or LAN gateways from offices,campuses, etc. using microwave radios. An exemplary backhaul connectionusing the microwave radios 132 is shown in FIG. 1. Traditionally, suchmicrowave radios 132 for backhaul have been mounted on high towers 112(or high rooftops of multi-story buildings) as shown in FIG. 1, suchthat each microwave radio 132 has an unobstructed line of sight (LOS)136 to the other. These microwave radios 132 can have data rates of 100Mb/s or higher at unobstructed LOS ranges of 300 m or longer withlatencies of 5 ms or less (to minimize overall network latency).

Traditional microwave backhaul radios 132 operate in a Point to Point(PTP) configuration using a single “high gain” (typically >30 dBi oreven >40 dBi) antenna at each end of the link 136, such as, for example,antennas constructed using a parabolic dish. Such high gain antennasmitigate the effects of unwanted multipath self-interference or unwantedco-channel interference from other radio systems such that high datarates, long range and low latency can be achieved. These high gainantennas however have narrow radiation patterns.

Furthermore, high gain antennas in traditional microwave backhaul radios132 require very precise, and usually manual, physical alignment oftheir narrow radiation patterns in order to achieve such highperformance results. Such alignment is almost impossible to maintainover extended periods of time unless the two radios have a clearunobstructed line of sight (LOS) between them over the entire range ofseparation. Furthermore, such precise alignment makes it impractical forany one such microwave backhaul radio to communicate effectively withmultiple other radios simultaneously (i.e., a “point to multipoint”(PMP) configuration).

In wireless edge access applications, such as cellular or WLAN, advancedprotocols, modulation, encoding and spatial processing across multipleradio antennas have enabled increased data rates and ranges for numeroussimultaneous users compared to analogous systems deployed 5 or 10 yearsago for obstructed LOS propagation environments where multipath andco-channel interference were present. In such systems, “low gain”(usually <6 dBi) antennas are generally used at one or both ends of theradio link both to advantageously exploit multipath signals in theobstructed LOS environment and allow operation in different physicalorientations as would be encountered with mobile devices. Althoughimpressive performance results have been achieved for edge access, suchresults are generally inadequate for emerging backhaul requirements ofdata rates of 100 Mb/s or higher, ranges of 300 m or longer inobstructed LOS conditions, and latencies of 5 ms or less.

In particular, “street level” deployment of cellular base stations, WLANaccess points or LAN gateways (e.g., deployment at street lamps, trafficlights, sides or rooftops of single or low-multiple story buildings)suffers from problems because there are significant obstructions for LOSin urban environments (e.g., tall buildings, or any environments wheretall trees or uneven topography are present).

FIG. 1 illustrates edge access using conventional unobstructed LOS PTPmicrowave radios 132. The scenario depicted in FIG. 1 is common for many2^(nd) Generation (2G) and 3^(rd) Generation (3G) cellular networkdeployments using “macrocells”. In FIG. 1, a Cellular Base TransceiverStation (BTS) 104 is shown housed within a small building 108 adjacentto a large tower 112. The cellular antennas 116 that communicate withvarious cellular subscriber devices 120 are mounted on the towers 112.The PTP microwave radios 132 are mounted on the towers 112 and areconnected to the BTSs 104 via an nT1 interface. As shown in FIG. 1 byline 136, the radios 132 require unobstructed LOS.

The BTS on the right 104 a has either an nT1 copper interface or anoptical fiber interface 124 to connect the BTS 104 a to the Base StationController (BSC) 128. The BSC 128 either is part of or communicates withthe core network of the cellular network operator. The BTS on the left104 b is identical to the BTS on the right 104 a in FIG. 1 except thatthe BTS on the left 104 b has no local wireline nT1 (or optical fiberequivalent) so the nT1 interface is instead connected to a conventionalPTP microwave radio 132 with unobstructed LOS to the tower on the right112 a. The nT1 interfaces for both BTSs 104 a, 104 b can then bebackhauled to the BSC 128 as shown in FIG. 1.

FIG. 2 is a block diagram of the major subsystems of a conventional PTPmicrowave radio 200 for the case of Time-Division Duplex (TDD)operation, and FIG. 3 is a block diagram of the major subsystems of aconventional PTP microwave radio 300 for the case of Frequency-DivisionDuplex (FDD) operation.

As shown in FIG. 2 and FIG. 3, the conventional PTP microwave radiotraditionally uses one or more (i.e. up to “n”) T1 interfaces 204 (or inEurope, E1 interfaces). These interfaces 204 are common in remote accesssystems such as 2G cellular base stations or enterprise voice and/ordata switches or edge routers. The T1 interfaces are typicallymultiplexed and buffered in a bridge (e.g., the Interface Bridge 208,308) that interfaces with a Media Access Controller (MAC) 212, 312.

The MAC 212, 312 is generally denoted as such in reference to asub-layer of Layer 2 within the Open Systems Interconnect (OSI)reference model. Major functions performed by the MAC include theframing, scheduling, prioritizing (or “classifying”), encrypting anderror checking of data sent from one such radio at FIG. 2 or FIG. 3 toanother such radio. The data sent from one radio to another is generallyin a “user plane” if it originates at the T1 interface(s) or in the“control plane” if it originates internally such as from the Radio LinkController (RLC) 248, 348 shown in FIG. 2 or FIG. 3. A typical MAC frameformat 400 (known as a MAC protocol data unit, or “MPDU”) with header404, frame body 408 and frame check sum (FCS) 412 is shown in FIG. 4.

With reference to FIGS. 2 and 3, the Modem 216, 316 typically resideswithin the “baseband” portion of the Physical (PHY) layer 1 of the OSIreference model. In conventional PTP radios, the baseband PHY, depictedby Modem 216, 316, typically implements scrambling, forward errorcorrection encoding, and modulation mapping for a single RF carrier inthe transmit path. In receive, the modem typically performs the inverseoperations of demodulation mapping, decoding and descrambling. Themodulation mapping is conventionally Quadrature Amplitude Modulation(QAM) implemented with In-phase (I) and Quadrature-phase (Q) branches.

The Radio Frequency (RF) 220, 320 also resides within the PHY layer ofthe radio. In conventional PTP radios, the RF 220, 320 typicallyincludes a single transmit chain (Tx) 224, 324 that includes I and Qdigital to analog converters (DACs), a vector modulator, optionalupconverters, a programmable gain amplifier, one or more channelfilters, and one or more combinations of a local oscillator (LO) and afrequency synthesizer. Similarly, the RF 220, 320 also typicallyincludes a single receive chain (Rx) 228, 328 that includes I and Qanalog to digital converters (ADCs), one or more combinations of an LOand a frequency synthesizer, one or more channel filters, optionaldownconverters, a vector demodulator and an automatic gain control (AGC)amplifier. Note that in many cases some of the one or more LO andfrequency synthesizer combinations can be shared between the Tx and Rxchains.

As shown in FIGS. 2 and 3, conventional PTP radios 200, 300 also includea single power amplifier (PA) 232, 332. The PA 232, 332 boosts thetransmit signal to a level appropriate for radiation from the antenna inkeeping with relevant regulatory restrictions and instantaneous linkconditions. Similarly, such conventional PTP radios 232, 332 typicallyalso include a single low-noise amplifier (LNA) 236, 336 as shown inFIGS. 2 and 3. The LNA 236, 336 boosts the received signal at theantenna while minimizing the effects of noise generated within theentire signal path.

As described above, FIG. 2 illustrates a conventional PTP radio 200 forthe case of TDD operation. As shown in FIG. 2, conventional PTP radios200 typically connect the antenna 240 to the PA 232 and LNA 236 via aband-select filter 244 and a single-pole, single-throw (SPST) switch242.

As described above, FIG. 3 illustrates a conventional PTP radio 300 forthe case of FDD operation. As shown in FIG. 3, in conventional PTPradios 300, then antenna 340 is typically connected to the PA 332 andLNA 336 via a duplexer filter 344. The duplexer filter 344 isessentially two band-select filters (tuned respectively to the Tx and Rxbands) connected at a common point.

In the conventional PTP radios shown in FIGS. 2 and 3, the antenna 240,340 is typically of very high gain such as can be achieved by aparabolic dish so that gains of typically >30 dBi (or even sometimes >40dBi), can be realized. Such an antenna usually has a narrow radiationpattern in both the elevation and azimuth directions. The use of such ahighly directive antenna in a conventional PTP radio link withunobstructed LOS propagation conditions ensures that the modem 216, 316has insignificant impairments at the receiver (antenna 240, 340) due tomultipath self-interference and further substantially reduces thelikelihood of unwanted co-channel interference due to other nearby radiolinks.

Although not explicitly shown in FIGS. 2 and 3, the conventional PTPradio may use a single antenna structure with dual antenna feedsarranged such that the two electromagnetic radiation patterns emanatedby such an antenna are nominally orthogonal to each other. An example ofthis arrangement is a parabolic dish. Such an arrangement is usuallycalled dual-polarized and can be achieved either by orthogonal verticaland horizontal polarizations or orthogonal left-hand circular andright-hand circular polarizations.

When duplicate modem blocks, RF blocks, and PA/LNA/switch blocks areprovided in a conventional PTP radio, then connecting each PHY chain toa respective polarization feed of the antenna allows theoretically up totwice the total amount of information to be communicated within a givenchannel bandwidth to the extent that cross-polarizationself-interference can be minimized or cancelled sufficiently. Such asystem is said to employ “dual-polarization” signaling.

When an additional circuit (not shown) is added to FIG. 2 that canprovide either the RF Tx signal or its anti-phase equivalent to eitherone or both of the two polarization feeds of such an antenna, then“cross-polarization” signaling can be used to effectively expand theconstellation of the modem within any given symbol rate or channelbandwidth. With two polarizations and the choice of RF signal or itsanti-phase, then an additional two information bits per symbol can becommunicated across the link. Theoretically, this can be extended andexpanded to additional phases, representing additional information bits.At the receiver, for example, a circuit (not shown) could detect if thetwo received polarizations are anti-phase with respect to each other, ornot, and then combine appropriately such that the demodulator in themodem block can determine the absolute phase and hence deduce the valuesof the two additional information bits. Cross-polarization signaling hasthe advantage over dual-polarization signaling in that it is generallyless sensitive to cross-polarization self-interference but for highorder constellations such as 64-QAM or 256-QAM, the relative increase inchannel efficiency is smaller.

In the conventional PTP radios shown in FIGS. 2 and 3, substantially allthe components are in use at all times when the radio link is operative.However, many of these components have programmable parameters that canbe controlled dynamically during link operation to optimize throughoutand reliability for a given set of potentially changing operatingconditions. The conventional PTP radios of FIGS. 2 and 3 control theselink parameters via a Radio Link Controller (RLC) 248, 348. The RLCfunctionality is also often described as a Link Adaptation Layer that istypically implemented as a software routine executed on amicrocontroller within the radio that can access the MAC 212, 312, Modem216, 316, RF 220, 320 and/or possibly other components with controllableparameters. The RLC 248, 348 typically can both vary parameters locallywithin its radio and communicate with a peer RLC at the other end of theconventional PTP radio link via “control frames” sent by the MAC 212,312 with an appropriate identifying field within a MAC Header 404 (inreference to FIG. 4).

Typical parameters controllable by the RLC 248, 348 for the Modem 216,316 of a conventional PTP radio include encoder type, encoding rate,constellation selection and reference symbol scheduling and proportionof any given PHY Protocol Data Unit (PPDU). Typical parameterscontrollable by the RLC 248, 348 for the RF 220, 320 of a conventionalPTP radio include channel frequency, channel bandwidth, and output powerlevel. To the extent that a conventional PTP radio employs twopolarization feeds within its single antenna, additional parameters mayalso be controlled by the RLC 248, 348 as self-evident from thedescription above.

In conventional PTP radios, the RLC 248, 348 decides, usuallyautonomously, to attempt such parameter changes for the link in responseto changing propagation environment characteristics such as, forexample, humidity, rain, snow, or co-channel interference. There areseveral well-known methods for determining that changes in thepropagation environment have occurred such as monitoring the receivesignal strength indicator (RSSI), the number of or relative rate of FCSfailures at the MAC 212, 312, and/or the relative value of certaindecoder accuracy metrics. When the RLC 248, 348 determines thatparameter changes should be attempted, it is necessary in most casesthat any changes at the transmitter end of the link become known to thereceiver end of the link in advance of any such changes. Forconventional PTP radios, and similarly for many other radios, there areat least two well-known techniques which in practice may not be mutuallyexclusive. First, the RLC 248, 348 may direct the PHY, usually in theModem 216, 316 relative to FIGS. 2 and 3, to pre-pend a PHY layerconvergence protocol (PLCP) header to a given PPDU that includes one ormore (or a fragment thereof) given MPDUs wherein such PLCP header hasinformation fields that notify the receiving end of the link ofparameters used at the transmitting end of the link. Second, the RLC248, 348 may direct the MAC 212, 312 to send a control frame, usually toa peer RLC 248, 348, including various information fields that denotethe link adaptation parameters either to be deployed or to be requestedor considered.

The foregoing describes at an overview level the typical structural andoperational features of conventional PTP radios which have been deployedin real-world conditions for many radio links where unobstructed (orsubstantially unobstructed) LOS propagation was possible. Theconventional PTP radio on a whole is completely unsuitable forobstructed LOS or PMP operation.

SUMMARY

The following summary of the invention is included in order to provide abasic understanding of some aspects and features of the invention. Thissummary is not an extensive overview of the invention and as such it isnot intended to particularly identify key or critical elements of theinvention or to delineate the scope of the invention. Its sole purposeis to present some concepts of the invention in a simplified form as aprelude to the more detailed description that is presented below.

Some embodiments of the claimed invention are directed to backhaulradios that are compact, light and low power for street level mounting,operate at 100 Mb/s or higher at ranges of 300 m or longer in obstructedLOS conditions with low latencies of 5 ms or less, can support PTP andPMP topologies, use radio spectrum resources efficiently and do notrequire precise physical antenna alignment. Radios with such exemplarycapabilities as described by multiple various embodiments are referredto herein by the term “Intelligent Backhaul Radio” (IBR). Exemplary IBRsare disclosed in detail in co-pending U.S. patent application Ser. No.13/212,036, entitled Intelligent Backhaul Radio, filed Aug. 17, 2011,and U.S. patent application Ser. No. 13/271,051, entitled IntelligentBackhaul System, filed Oct. 11, 2011, U.S. patent application Ser. No.13/415,778, entitled Intelligent Backhaul System, filed Mar. 8, 2012,U.S. patent application Ser. No. 13/448,294, entitled Hybrid BandIntelligent Backhaul Radio, filed Apr. 16, 2012, U.S. patent applicationSer. No. 13/371,366, entitled Intelligent Backhaul Radio, filed Feb. 10,2012, the entireties of which is hereby incorporated by reference.

The IBR provides reliable, high-capacity data communications in urbandeployments. In some deployments, there is not line-of-sight visibilitybetween the two end nodes of the link, and in such cases the propagationchannel relies on multipath scattering and diffraction to achievecommunication. The resulting wireless channel features high path lossexponents, time-domain delay spread, and wide angular-spread of incomingpower (i.e spatial Rayleigh fading). In addition, some embodiments ofthe IBR rely on two-stream spatial multiplexing, which places furtherdemands on the antenna design.

To achieve the desired throughput and reliability in the aboveconditions and utilizing specific embodiments, the antenna system mustmeet the following specifications:

-   -   Wide azimuthal coverage (e.g. 100 to 120 degrees) to accommodate        for the wide-angular spread of incoming power;    -   Narrow vertical beamwidths (e.g. 15 degrees);    -   At minimum two uncorrelated antenna ports for any direction        within the coverage zone;    -   High-gain to accommodate the high-path loss; and    -   Compact overall enclosure.

Fundamentally, there are three diversity mechanisms available to achieveuncorrelated signals at the terminals of an antenna: spatial diversity,polarization diversity, and angular beam diversity. The optimalselection or combination of these diversity mechanisms depends on apriori knowledge of typical channel characteristics.

For example, in a wide angle-of-arrival environment, uncorrelatedsignals can be achieved with omni-directional antennas with relativelycompact spatial separation (on the order of one wavelength) e.g. mostconsumer-grade WiFi access points utilize spatial diversity, as known inthe art. This spacing requirement increases as the channel becomes moredirectional, making spatial diversity a poor choice for directionalscenarios when minimal physical size is important.

Polarization diversity has the advantage that a well-designeddual-polarity antenna can be achieved in virtually the same overalldimensions as the equivalent single-polarization antenna. Polarizationdiversity is advantageous where there is a channel with significantpolarization-dependent scattering. Polarization diversity typicallyrequires that the polarizations be orthogonal, i.e. the relative anglebetween the two antennas should be 90 degrees; however, the twopolarizations can be arbitrarily rotated with regards to the earth.

Angular beam diversity is particularly effective if the incoming angularpower distribution is clustered around more than one dominant angles ofarrival. The angle between these clusters dictates the minimum angularwidth of the antenna pattern that will achieve uncorrelated antennas.Angular beam diversity has two advantages over spatial diversity, gainand compactness. As the beamwidth of an antenna pattern narrows the gainof the antenna increases, and the sources for each beam can beco-located without any loss in diversity. For example, in analogbeamforming schemes such as a butler matrix, the same antenna elementsare re-used to form the various directional beams.

According to an aspect of the invention, an intelligent backhaul radiois disclosed that includes a directional transmit antenna array, forsimultaneously transmitting a plurality of transmit symbol streams to atarget radio, said directional transmit antenna array comprising aplurality of transmit antenna sub-arrays, wherein each of said pluralityof transmit antenna sub-arrays comprise a plurality of transmit antennaelements; a plurality of directional receive antenna arrays, forsimultaneously receiving a plurality of receive RF signals comprising aplurality of receive symbol streams from the target radio, wherein eachof said plurality of directional receive antenna arrays comprise aplurality of receive antenna sub-arrays, wherein each of said pluralityof receive antenna sub-arrays comprise a plurality of receive antennaelements, wherein each of said receive antenna sub-arrays has anormalized antenna pattern similar to the normalized antenna pattern ofeach of the transmit antenna sub-arrays in elevation and narrower thanthe normalized antenna pattern of each of the transmit antennasub-arrays in azimuth, and wherein each of the plurality of directionalreceive antenna arrays are positioned physically offset from saiddirectional transmit antenna array, and distributed in azimuthalorientation such that the aggregate normalized azimuthal antennapatterns of the plurality of receive antenna sub-arrays in composite isequal to or greater than the normalized antenna pattern of each of thetransmit antenna sub-arrays in azimuth; one or more demodulator cores,wherein each demodulator core demodulates one or more of said pluralityof receive symbol streams to produce a respective receive data interfacestream; a plurality of receive RF chains to convert the plurality ofreceive RF signals to a plurality of respective receive chain outputsignals; a frequency selective receive path channel multiplexer betweenthe one or more demodulator cores and the plurality of receive RFchains, the frequency selective receive path channel multiplexer togenerate the one or more receive symbol streams from the plurality ofreceive chain output signals; and one or more selectable RF connectionsthat selectively couple certain of the plurality of said receive antennasub-arrays to certain of the plurality of receive RF chains, wherein thenumber of receive antenna sub-arrays that can be selectively coupled toreceive RF chains exceeds the number of receive RF chains that canaccept receive RF signals from the one or more selectable RFconnections; and a radio resource controller, wherein the radio resourcecontroller sets or causes to be set the specific selective couplingsbetween the certain of the plurality of receive antenna sub-arrays andthe certain of the plurality of receive RF chains.

Each of said plurality of transmit sub-arrays may transmit one or moreof said plurality of transmit symbol streams, and at least two of saidplurality of transmit sub-arrays may utilize differing polarizations.

The plurality of transmit antenna elements may be printed dipole antennaelements and said plurality of receive antenna elements may be patchantenna elements.

One or more of said receive patch antenna elements may be utilized by aplurality of the receive antenna sub-arrays of the directional receiveantenna array, said receive patch antenna element utilizing separateantenna feed structures providing for substantially orthogonallypolarized reception for each of the sub-arrays.

At least one of said one or more of said plurality of transmit symbolstreams may be multiplexed into a plurality of transmit RF signals. Theplurality of transmit RF signals may be respectively transmitted fromeach of said plurality of transmit antenna sub-arrays. At least two ofsaid RF signals may include the same transmit symbol stream.

The same transmit symbol stream may be modified in gain, phase, or delayprior to transmission by each of said plurality of transmit sub arraysto achieve a composite transmit symbol stream property.

The composite transmit symbol stream property may be a modifiedpolarization.

The composite transmit symbol stream property may result in an improvedreceive property at a receiver of the target radio.

The improved receive property may be one or more of: a received RSSI, areceived signal to noise ratio, a signal to interference ratio, ade-correlation between differing transmit streams, a propagation channelproperty, a delay spread, and a signal variability.

A metric may be provided by said target radio to the intelligentbackhaul radio, said metric used to adapt said gain, phase, or delay ofsaid transmit symbol stream prior to transmission by each of saidplurality of transmit sub arrays, wherein said metric is derived fromthe receive property at the target radio.

At least one of said plurality of transmit symbol streams may include aunique transmit symbol stream.

The plurality of transmit antenna sub-arrays of said directionaltransmit antenna array may utilize a common ground plane, and saidground plane may include non-conducting slots at the edge of said groundplane. The slots may be aligned with said printed dipole antennaelements of a first of the transmit sub-arrays to effect an associatedantenna pattern of that sub-array, and substantially not affect theantenna pattern of a second transmit sub-array of said plurality oftransmit sub-arrays utilizing said shared ground plane. The secondtransmit sub-array may include antenna elements having a polarizationorthogonal to the polarization of said first transmit sub-array.

The plurality of receive antenna arrays positioned physically offsetfrom said directional transmit antenna array may be distributedlinearly. The linear distribution of said plurality of directionalreceive antenna arrays may be vertical. The linear distribution of saidplurality of directional receive antenna arrays may be horizontal.

The plurality of transmit antenna sub-arrays of said directionaltransmit antenna array may utilize a common ground plane and each ofsaid printed dipole antenna elements of a first transmit sub-array ofsaid plurality of transmit sub-arrays may utilize a first polarizationof said at least two differing polarizations, and may be printed on asingle printed circuit board.

Each of said printed dipole antenna elements of a second sub-array ofsaid plurality of transmit sub-arrays may utilize a second polarizationof said at least two differing polarizations, and may be printed onindividual printed circuit boards.

The single printed circuit board of said first sub-array may includeslots for mechanically aligning, securing, or joining with complimentaryslots located within said individual printed circuit boards of saidsecond sub-array.

Each of the individual printed circuit boards and the single printedcircuit board may include tabs for mating with slots in a printedcircuit board of said common ground plane and for mechanically securingsaid printed dipole antenna elements.

Each of the individual printed circuit boards and the single printedcircuit board may include tabs for mating with slots in a printedcircuit board of said common ground plane and for coupling feed networksdisposed upon said printed circuit board of said common ground planewith said printed dipole antenna elements.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated into and constitute apart of this specification, illustrate one or more examples ofembodiments and, together with the description of example embodiments,serve to explain the principles and implementations of the embodiments.

FIG. 1 is an illustration of conventional point to point (PTP) radiosdeployed for cellular base station backhaul with unobstructed line ofsight (LOS).

FIG. 2 is a block diagram of a conventional PTP radio for Time DivisionDuplex (TDD).

FIG. 3 is a block diagram of a conventional PTP radio for FrequencyDivision Duplex (FDD).

FIG. 4 is an illustration of a MAC Protocol Data Unit (MPDU).

FIG. 5 is an illustration of intelligent backhaul radios (IBRs) deployedfor cellular base station backhaul with obstructed LOS according to oneembodiment of the invention.

FIG. 6 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 7 is a block diagram of an IBR according to one embodiment of theinvention.

FIG. 8 is a block diagram illustrating an exemplary deployment of IBRsaccording to one embodiment of the invention.

FIG. 9 is a block diagram illustrating an exemplary deployment of IBRsaccording to one embodiment of the invention.

FIG. 10 is a block diagram of an IBR antenna array according to oneembodiment of the invention.

FIG. 10A is a block diagram of an IBR antenna array according to oneembodiment of the invention.

FIG. 11 is a block diagram of a front-end unit for TDD operationaccording to one embodiment of the invention.

FIG. 12 is a block diagram of a front-end unit for FDD operationaccording to one embodiment of the invention.

FIG. 12A is a block diagram of a front-end transmission unit accordingto one embodiment of the invention.

FIG. 12B is a block diagram of a front-end reception unit according toone embodiment of the invention.

FIG. 13 is a perspective view of an IBR according to one embodiment ofthe invention.

FIG. 14 is a perspective view of an IBR according to one embodiment ofthe invention.

FIG. 15 is a perspective view of an IBR according to one embodiment ofthe invention.

FIG. 16 is a block diagram illustrating an exemplary transmit chainwithin an IBR RF according to one embodiment of the invention.

FIG. 17 is a block diagram illustrating an exemplary receive chainwithin an IBR RF according to one embodiment of the invention.

FIG. 18 is a block diagram illustrating an IBR modem according to oneembodiment of the invention.

FIG. 19 is a block diagram illustrating a modulator core j according toone embodiment of the invention.

FIG. 20 is a block diagram illustrating a demodulator core j accordingto one embodiment of the invention.

FIG. 21 is a block diagram illustrating a modulator core j according toone embodiment of the invention.

FIG. 22 is a block diagram illustrating a demodulator core j accordingto one embodiment of the invention.

FIG. 23, consisting of FIG. 23A and FIG. 23B, is a block diagramillustrating a channel multiplexer (MUX) according to one embodiment ofthe invention. FIG. 23A is a partial view showing the transmit path andthe channel equalizer coefficients generator within the exemplarychannel MUX. FIG. 23B is a partial view showing the receive path withinthe exemplary channel MUX.

FIG. 24 is a block diagram illustrating an exemplary Tx-CE-m accordingto one embodiment of the invention.

FIG. 24A is a block diagram illustrating an exemplary Tx-CE-m, includinga complex circular FIR capability, according to one embodiment of theinvention.

FIG. 24B is a plot of an exemplary desired transmit equalizer impulseresponse for use with the Tx-CE-m of FIG. 24A according to oneembodiment of the invention.

FIG. 24C is a plot of an exemplary desired transmit equalizer frequencyresponse for use with the Tx-CE-m of FIG. 24A according to oneembodiment of the invention.

FIG. 24D is a plot of an exemplary desired transmit equalizer impulseresponse for use with the Tx-CE-m of FIG. 24A according to oneembodiment of the invention.

FIG. 24E is a plot of an exemplary 16 QAM—10 symbol Transmit Block({right arrow over (TxBlock)}_(k)) for processing with the Tx-CE-m ofFIG. 24A according to one embodiment of the invention.

FIG. 24F is a plot of the result of a linear convolution of theexemplary 16 QAM—10 symbol Transmit Block ({right arrow over(TxBlock)}_(k)) according to one embodiment of the invention.

FIG. 24G is a plot of the result of the FIR based circular convolutionaccording to one embodiment of the invention.

FIG. 24H is a plot of the result of a frequency domain based circularconvolution of the exemplary 16 QAM—10 symbol Transmit Block ({rightarrow over (TxBlock)}_(k)) according to one embodiment of the invention.

FIG. 24I is a magnitude plot of the result of both the circular FIR andthe frequency domain based circular convolutions of the exemplary 16QAM—10 symbol Transmit Block ({right arrow over (TxBlock)}_(k))according to one embodiment of the invention.

FIG. 25 is a block diagram illustrating an exemplary Rx-CE-l accordingto one embodiment of the invention.

FIG. 26 is a timing diagram illustrating processing of PPDU-l withTx-path and Rx-path of respective IBR channel MUXs according to oneembodiment of the invention.

FIG. 27 is a block diagram illustrating an exemplary Tx PLCP accordingto one embodiment of the invention.

FIG. 28 is a block diagram illustrating an exemplary Tx PLCP Mod-jaccording to one embodiment of the invention.

FIG. 29 is a block diagram illustrating an exemplary Rx PLCP accordingto one embodiment of the invention.

FIG. 30 is a block diagram illustrating an exemplary Rx PLCP Mod-jaccording to one embodiment of the invention.

FIG. 31 is a schematic diagram of an IBR communications protocols stackaccording to one embodiment of the invention.

FIG. 32 is a schematic diagram of an IBR communications protocols stackaccording to one embodiment of the invention.

FIG. 33 is a block diagram of an IBR media access control (MAC)according to one embodiment of the invention.

FIG. 34 is a timing diagram illustrating channel activity for FDD withfixed superframe timing according to one embodiment of the invention.

FIG. 35 is a timing diagram illustrating channel activity for TDD withfixed superframe timing according to one embodiment of the invention.

FIG. 36 is a timing diagram illustrating channel activity for TDD/CSMAwith variable superframe timing according to one embodiment of theinvention.

FIG. 37 is a diagram illustrating a sector antenna panel and processingaccording to one embodiment of the invention.

FIG. 38 is a plot of an exemplary far field antenna pattern of a sectorantenna panel according to one embodiment of the invention.

FIG. 39 is a diagram illustrating an alternative sector antenna paneland processing, including separate transmit and receive antennasaccording to one embodiment of the invention.

FIG. 40 is a plot of an exemplary far field antenna pattern of analternative sector antenna panel including separate transmit and receiveantennas according to one embodiment of the invention.

FIG. 41 is a diagram illustrating an alternative sector antenna paneland processing including separate transmit and receive antennasaccording to one embodiment of the invention.

FIG. 42 is a plot of an exemplary far field antenna pattern of analternative sector antenna panel and processing including separatetransmit and receive antennas according to one embodiment of theinvention.

FIG. 43 is a diagram of a dual-polarity, two-port patch antenna elementincluding feed and grounding points according to one embodiment of theinvention.

FIG. 44A is a diagram of a front view of an exemplary dual-polarity, twoport, patch antenna array according to one embodiment of the invention.

FIG. 44B is a diagram of a side view of an exemplary dual-polarity, twoport, patch antenna array according to one embodiment of the invention

FIG. 45A is a diagram of a view of an exemplary single-polarity, singleport, printed dipole antenna element for use in an antenna arrayaccording to one embodiment of the invention.

FIG. 45B is a diagram of an alternative view of an exemplarysingle-polarity, single port, printed dipole antenna element for use inan antenna array according to one embodiment of the invention

FIG. 46A is a diagram of a view of an exemplary dual-polarity, two port,antenna array utilizing printed dipole antenna elements according to oneembodiment of the invention.

FIG. 46B is a diagram of a front view of an exemplary dual-polarity, twoport, antenna array utilizing printed dipole antenna elements accordingto one embodiment of the invention.

FIG. 46C is a diagram of a back view of an exemplary dual-polarity, twoport, antenna array utilizing printed dipole antenna elements, showing acooperate feed network for each single polarized sub-array associatedwith each port according to one embodiment of the invention.

FIG. 47A is a plot of an exemplary far field elevation antenna patternof a dual-polarity, two port, antenna array utilizing printed dipoleantenna elements according to one embodiment of the invention.

FIG. 47B is a plot of an exemplary far field azimuthal antenna patternof a dual-polarity, two port, antenna array utilizing printed dipoleantenna elements according to one embodiment of the invention.

FIG. 48 is a diagram of a view of an alternative exemplarydual-polarity, two port, antenna array utilizing printed dipole antennaelements, including alternative printed dipole antenna elements andfurther providing vertical and horizontal polarizations according to oneembodiment of the invention.

FIG. 49A is a diagram of an alternative exemplary printed dipole antennaelement according to one embodiment of the invention.

FIG. 49B is a diagram of an alternative exemplary printed dipole antennaincluding illustrative ground plane interface according to oneembodiment of the invention.

FIG. 50 is a diagram of an exemplary printed dipole antenna structurefor use in an antenna array according to one embodiment of theinvention.

FIG. 51 is a diagram depicting an exemplary assembly a dual-polarized,two port, printed dipole antenna array utilizing a printed dipolestructure according to one embodiment of the invention.

FIG. 52A is a diagram of a front view of an exemplary horizontallyarranged intelligent backhaul radio antenna array according to oneembodiment of the invention.

FIG. 52B is a diagram of an alternative view of an exemplaryhorizontally arranged intelligent backhaul radio antenna array accordingto one embodiment of the invention.

FIG. 52C is a diagram of a top of an exemplary horizontally arrangedintelligent backhaul radio antenna array according to one embodiment ofthe invention.

FIG. 53A is a diagram of a front view of an exemplary verticallyarranged intelligent backhaul radio antenna array according to oneembodiment of the invention.

FIG. 53B is a diagram of an alternative view of an exemplary verticallyarranged intelligent backhaul radio antenna array according to oneembodiment of the invention.

FIG. 53C is a diagram of a side view of an exemplary vertically arrangedintelligent backhaul radio antenna array according to one embodiment ofthe invention.

FIG. 53D is a diagram of a top view of an exemplary vertically arrangedintelligent backhaul radio antenna array according to one embodiment ofthe invention.

DETAILED DESCRIPTION

FIG. 5 illustrates deployment of intelligent backhaul radios (IBRs) inaccordance with an embodiment of the invention. As shown in FIG. 5, theIBRs 500 are deployable at street level with obstructions such as trees504, hills 508, buildings 512, etc. between them. The IBRs 500 are alsodeployable in configurations that include point to multipoint (PMP), asshown in FIG. 5, as well as point to point (PTP). In other words, eachIBR 500 may communicate with more than one other IBR 500.

For 3G and especially for 4^(th) Generation (4G), cellular networkinfrastructure is more commonly deployed using “microcells” or“picocells.” In this cellular network infrastructure, compact basestations (eNodeBs) 516 are situated outdoors at street level. When sucheNodeBs 516 are unable to connect locally to optical fiber or a copperwireline of sufficient data bandwidth, then a wireless connection to afiber “point of presence” (POP) requires obstructed LOS capabilities, asdescribed herein.

For example, as shown in FIG. 5, the IBRs 500 include an Aggregation EndIBR (AE-IBR) and Remote End IBRs (RE-IBRs). The eNodeB 516 of the AE-IBRis typically connected locally to the core network via a fiber POP 520.The RE-IBRs and their associated eNodeBs 516 are typically not connectedto the core network via a wireline connection; instead, the RE-IBRs arewirelessly connected to the core network via the AE-IBR. As shown inFIG. 5, the wireless connection between the IBRs include obstructions(i.e., there may be an obstructed LOS connection between the RE-IBRs andthe AE-IBR).

FIGS. 6 and 7 illustrate exemplary embodiments of the IBRs 500 shown inFIG. 5. In FIGS. 6 and 7, the IBRs 500 include interfaces 604, interfacebridge 608, MAC 612, modem 624, channel MUX 628, RF 632, which includesTx1 . . . TxM 636 and Rx1 . . . RxN 640, antenna array 648 (includesmultiple antennas 652), a Radio Link Controller (RLC) 656 and a RadioResource Controller (RRC) 660. The IBR may optionally include an IBMSagent 700 as shown in FIG. 7. It will be appreciated that the componentsand elements of the IBRs may vary from that illustrated in FIGS. 6 and7. The various elements of the IBR are described below by theirstructural and operational features in numerous different embodiments.

The external interfaces of the IBR (i.e., the IBR Interface Bridge 608on the wireline side and the IBR Antenna Array 648 (including antennas652) on the wireless side) are a starting point for describing somefundamental differences between the numerous different embodiments ofthe IBR 500 and the conventional PTP radios described above (or othercommonly known radio systems, such as those built to existing standardsincluding 802.11n (WiFi) and 802.16e (WiMax)).

In some embodiments, the IBR Interface Bridge 608 physically interfacesto standards-based wired data networking interfaces 604 as Ethernet 1through Ethernet P. “P” represents a number of separate Ethernetinterfaces over twisted-pair, coax or optical fiber. The IBR InterfaceBridge 608 can multiplex and buffer the P Ethernet interfaces 604 withthe IBR MAC 612. For the case of P=1 (a single Ethernet interface), themultiplexing operation of the IBR Interface Bridge 608 is a trivial“pass-through” between the single Ethernet interface and the buffer. Inexemplary embodiments, the IBR Interface Bridge 608 preserves “Qualityof Service” (QoS) or “Class of Service” (CoS) prioritization asindicated, for example, in IEEE 802.1q 3-bit Priority Code Point (PCP)fields within the Ethernet frame headers, such that either the IBR MAC612 schedules such frames for transmission according to policiesconfigured within the IBR of FIG. 6 or communicated via the IBMS Agent700 of FIG. 7, or the IBR interface bridge 608 schedules the transfer ofsuch frames to the IBR MAC 612 such that the same net effect occurs. Inother embodiments, the IBR interface bridge 608 also forwards andprioritizes the delivery of frames to or from another IBR over aninstant radio link based on Multiprotocol Label Switching (MPLS) orMultiprotocol Label Switching Transport Profile (MPLS-TP).

In some embodiments, the IBR Interface Bridge 608 can also perform layer2 switching of certain Ethernet interfaces to other Ethernet interfaces604 in response to radio link failure conditions and policies configuredwithin the IBR of FIG. 6 or communicated via the IBMS Agent 700 of FIG.7. A radio link failure condition can arise from any one or more ofmultiple possible events, such as the following exemplary radio linkfailure condition events:

-   -   physical failure of a component within the IBR other than the        IBR Interface Bridge and its power supply;    -   degradation of the RF link beyond some pre-determined throughput        level due to either changing propagation environment or        additional co-channel interference; and    -   failure of any kind at the other end of the RF link that        prevents connection to the ultimate source or destination.

FIG. 8 illustrates an exemplary configuration of multiple IBRs (IBR 1 1804, IBR 2 808). Each IBR 804, 808 has layer 2 switching in the IBRinterface bridges 816, 820 (each corresponding to an instance of the IBRInterface Bridge 608 of FIGS. 6 and 7). In one embodiment, the datanetworking traffic can be from, for example, a cellular eNodeB, a WiFihotspot, a enterprise edge router, or any of numerous other remote datanetworks 828. As shown in FIG. 8, the remote data network 828 isconnected via an Ethernet cable (copper or fiber) 832 to the firstEthernet port 836 on the IBR Interface Bridge 816 of IBR1 804. AnotherEthernet cable 840 connects the second Ethernet port 844 on the IBRInterface Bridge 816 of IBR1 804 to the first Ethernet port 848 on theIBR Interface Bridge 820 of IBR2 808. If a radio link failure conditionoccurs for any reason, such as those listed above, with respect to RFLink 1 852, then the layer 2 switch within the IBR Interface Bridge 816of IBR1 804 can automatically connect all data networking trafficoriginating from or destined to Remote Data Network 1 828 via IBR2 808and RF Link2 856, completely transparently to Remote Data Network 1 828.This provides fail over redundancy to reduce the probability of networkoutage at Remote Data Network 1 828.

In some embodiments, the IBR Interface Bridge with layer 2 switching canalso be configured to perform load balancing in response to operatingconditions and policies configured within the IBR of FIG. 6 orcommunicated via the IBMS Agent 700 of FIG. 7. For example, withreference to FIG. 8, the layer 2 switch in the IBR Interface Bridge 816of IBR1 804 can connect all data networking traffic in excess of datarates above a certain limit, such as a pre-determined level or theinstantaneous supportable rate on RF Link 1 852, over to IBR2 808. Forfull two-way functionality of this load balancing, an analogous loadbalancing capability exists within the layer 2 switch of an IBR at therespective other ends of both RF Link 1 852 and RF Link 2 856.

FIG. 9 illustrates an alternative configuration of IBRs 804, 808 withlayer 2 switching capability and optional load balancing capability forthe case of two disparate Remote Data Networks 1 and 2 (900 and 904,respectively). The two Remote Data Networks 1 and 2 (900 and 904) are ator within Ethernet cabling distance of two IBRs 1 and 2 (804, 808)operating on two RF links 1 and 2 (852, 856). As described above, theRemote Data Network 828 is connected via Ethernet cable 832 to the firstEthernet port 836, and the IBRs 1 and 2 (804, 808) are connected viaEthernet cable 840 at Ethernet ports 844 and 848 respectively. TheRemote Data Network 2 904 is connected to IBR 2 808 via Ethernet cable908 at the Ethernet port 912. In the embodiment shown in FIG. 9, if aradio link failure condition occurs for any reason, such as those listedabove, with respect to RF Link 1 852, then the IBR Interface Bridge 816of IBR1 804 can use its layer 2 switch to connect to the IBR InterfaceBridge 820 of IBR2 808 via Ethernet cable 840 such that IBR2 808 canbackhaul, subject to its throughput capabilities and prioritizationpolicies, the traffic originating from or destined to both Remote DataNetwork 1 900 and Remote Data Network 2 904. Similarly, the IBRs 804,808 can perform load balancing across both RF Links 1 and 2 (852, 856),for traffic originating from or destined to Remote Data Networks 1 and 2(900, 904).

In some embodiments, RF link 1 852 may utilize spectrum possessingadvantageous conditions, such as reduced interference, wider channelbandwidth, better propagation characteristics, and/or higher allowablepower than the spectrum utilized by RF Link 2 856, or vice versa. In thesituation where a radio link failure condition occurs with respect tothe more advantageous spectrum, either control signaling between the twoIBR Interface Bridges 816, 820 of IBRs 1 and 2 as shown in FIG. 6 ormessaging between the two IBMS Agents 700 as shown in FIG. 7, whetherdirectly or indirectly via one or more intermediaries, can cause theredundant IBR 808 to change to the advantageous spectrum no longer beingused by RF Link 852 with the failure condition.

FIG. 10 illustrates an exemplary embodiment of an IBR Antenna Array 648.FIG. 10 illustrates an antenna array having Q directive gain antennas652 (i.e., where the number of antennas is greater than 1). In FIG. 10,the IBR Antenna Array 648 includes an IBR RF Switch Fabric 1012, RFinterconnections 1004, a set of Front-ends 1008 and the directive gainantennas 652. The RF interconnections 1004 can be, for example, circuitboard traces and/or coaxial cables. The RF interconnections 1004 connectthe IBR RF Switch Fabric 1012 and the set of Front-ends 1008. EachFront-end 1008 is associated with an individual directive gain antenna652, numbered consecutively from 1 to Q.

FIG. 10A illustrates an additional exemplary embodiment of an IBRAntenna Array 648, and comprises a block diagram of an IBR antenna arrayaccording to one embodiment of the invention relating to the use ofdedicated transmission and reception antennas. In some IBR embodimentsthe embodiment of FIG. 10 may be replaced with the embodiments describedin relation to FIG. 10A. For instance, such substitution may be made inuse with either FDD, TDD, or even non-conventional duplexing systems.FIG. 10A illustrates an antenna array having Q_(R)+Q_(T) directive gainantennas 652 (i.e., where the number of antennas is greater than 1). InFIG. 10A, the IBR Antenna Array 648 includes an IBR RF Switch Fabric1012, RF interconnections 1004, a set of Front-ends 1009 and 1010 andthe directive gain antennas 652. The RF interconnections 1004 can be,for example, circuit board traces and/or coaxial cables. The RFinterconnections 1004 connect the IBR RF Switch Fabric 1012 and the setof Front-end Transmission Units 1009 and the set of Front-end ReceptionUnits 1010. Each Front-end transmission unit 1009 is associated with anindividual directive gain antenna 652, numbered consecutively from 1 toQ_(T). Each Front-end reception unit 1010 is associated with anindividual directive gain antenna 652, numbered consecutively from 1 toQ_(R). The present embodiment may be used, for example, with the antennaarray embodiments of FIG. 39, FIG. 41, FIG. 52, FIG. 53, or embodimentsdescribed elsewhere. Such dedicated transmission antennas are coupled tofront-end transmission units 1009 and comprise antenna element 652.

In alternative embodiment, the IBR RF Switch fabric 1012 may be bypassedfor the transmission signals when the number of dedicated transmissionantennas and associated front-end transmission units (Q_(T)) is equal tothe number of RF transmission signals RF-Tx-M (e.g. Q_(T)=M), resultingin directly coupling the IBR RF 632 transmissions to respectivetransmission front-end transmission units 1009. The dedicated receptionantennas, comprising an antenna element 652 in some embodiments, arecoupled to front-end reception units 1010, which in the presentembodiment are coupled to the IBR RF Switch Fabric. In an additionalalternative embodiment, the IBR RF Switch fabric 1012 may be bypassedfor the reception signals when the number of dedicated receptionantennas and associated front-end reception units (Q_(R)) is equal tothe number of RF reception signals RF-Rx-N (e.g. Q_(R)=N), resulting indirectly coupling the IBR RF 632 reception ports to respective front-endreception units 1010.

FIG. 11 illustrates an exemplary embodiment of the Front-end circuit1008 of the IBR Antenna Array 648 of FIG. 10 for the case of TDDoperation, and FIG. 12 illustrates an exemplary embodiment of theFront-end circuit 1008 of the IBR Antenna Array 648 of FIG. 10 for thecase of FDD operation. The Front-end circuit 1008 of FIG. 11 includes atransmit power amplifier PA 1104, a receive low noise amplifier LNA1108, SPDT switch 1112 and band-select filter 1116. The Front-endcircuit 1008 of FIG. 12 includes a transmit power amplifier PA 1204,receive low noise amplifier LNA 1208, and duplexer filter 1212. Thesecomponents of the Front-end circuit are substantially conventionalcomponents available in different form factors and performancecapabilities from multiple commercial vendors.

As shown in FIGS. 11 and 12, each Front-end 1008 also includes an“Enable” input 1120, 1220 that causes substantially all active circuitryto power-down. Power-down techniques are well known. Power-down isadvantageous for IBRs in which not all of the antennas are utilized atall times. It will be appreciated that alternative embodiments of theIBR Antenna Array may not utilize the “Enable” input 1120, 1220 orpower-down feature. Furthermore, for embodiments with antenna arrayswhere some antenna elements are used only for transmit or only forreceive, then certain Front-ends (not shown) may include only thetransmit or only the receive paths of FIGS. 11 and 12 as appropriate.

FIG. 12A is a block diagram of a front-end transmission unit accordingto one embodiment of the invention relating to the use of dedicatedtransmission and reception antennas, and FIG. 12B is a block diagram ofa front-end reception unit according to one embodiment of the inventionrelating to the use of dedicated transmission and reception antennas. Asshown in FIGS. 12A and 12B, each Front-end 1008 also includes an“Enable” input 1225, 1230 that causes substantially all active circuitryto power-down, and any known power-down technique may be used.Power-down is advantageous for IBRs in which not all of the antennas areutilized at all times. It will be appreciated that alternativeembodiments of the IBR Antenna Array may not utilize the “Enable” input1225, 1230 or power-down feature. Furthermore, for some embodimentsassociated with FIG. 10A for example (with antenna arrays where someantenna elements are used only for transmit or only for receive) thencertain Front-ends may include only the transmit 1109 or only thereceive paths 1010 of FIGS. 12A and 12B as appropriate. With respect toFIG. 12A, Bandpass filter 1240 receives transmission signal RF-SW-Tx-qt,provides filtering and couples the signal to power amplifier 1104, thento low pass filter 1050. The output of the lowpass filter is thencoupled to dedicated transmission antenna, which is comprised ofdirective antenna element 652. With respect to FIG. 12B, directiveantenna element 652 is a dedicated receive only antenna and coupled toreceive filter 1270, when is in turn coupled to LNA 1208. The resultingamplified receive signal is coupled to band bass filter 1260, whichprovides output RF-SW-Rx-qr.

As described above, each Front-end (FE-q) corresponds to a particulardirective gain antenna 652. Each antenna 652 has a directivity gain Gq.For IBRs intended for fixed location street-level deployment withobstructed LOS between IBRs, whether in PTP or PMP configurations, eachdirective gain antenna 652 may use only moderate directivity compared toantennas in conventional PTP systems at a comparable RF transmissionfrequency. Based on measurements of path loss taken at street level at2480 MHz in various locations in and nearby San Jose, Calif. duringAugust and September 2010, IBR antennas should have a Gq of at least 6dBi, and, in typical embodiments for operation between 2 GHz and 6 GHzRF transmission frequency, a Gq in the range of 10-18 dBi, wherein theradiation pattern in the elevation direction is typically less than 90°and nominally parallel to the local surface grade. At higher RFtransmission frequencies, higher gain ranges of Gq are expected to bepreferable. For example, Gq may be preferably 16-24 dBi for 20-40 GHzoperation or 20-28 dBi for 60-90 GHz operation. In one particularembodiment, the directive gain antennas 652 are “patch” type antennaswith Gq of about 13 dBi and nominally equal elevation and azimuthradiation patterns (about 40° each). Patch type antennas areadvantageous because they can be realized using substantiallyconventional printed circuit board (PCB) fabrication techniques, whichlowers cost and facilitates integration with Front-end circuitcomponents and/or some or substantially all of the IBR RF Switch Fabric.However, may other antenna types, such as helical, horn, and slot, aswell as microstrip antennas other than patch (such as reflecteddipoles), and the like, may be used with the IBR Antenna Array. In analternative embodiment, the directive gain antennas 652 are reflecteddipoles with Gq of about 15 dBi (about 50° azimuth and 20° elevation).In many embodiments, the antenna elements are chosen with elevationangular patterns considerably less than azimuthal angular patterns.

In the IBR Antenna Array 648 illustrated in FIGS. 6, 7 and 10, the totalnumber of individual antenna elements 652, Q, is greater than or equalto the larger of the number of RF transmit chains 636, M, and the numberof RF receive chains 640, N. In some embodiments, some or all of theantennas 652 may be split into pairs of polarization diverse antennaelements realized by either two separate feeds to a nominally singleradiating element or by a pair of separate orthogonally orientedradiating elements. Such cross polarization antenna pairs enable eitherincreased channel efficiency or enhanced signal diversity as describedfor the conventional PTP radio. The cross-polarization antenna pairs aswell as any non-polarized antennas are also spatially diverse withrespect to each other.

In some embodiments, certain antenna elements 652 may be configured withdifferent antenna gain Gq and/or radiation patterns compared to othersin the same IBR to provide pattern diversity.

In some embodiments, some antenna elements 652 may be oriented indifferent ways relative to others to achieve directional diversity. Forexample, FIG. 13 illustrates an IBR suitable for obstructed LOS PTPoperation (or sector-limited PMP operation) in which spatial diversity(and optionally polarization diversity and/or pattern diversity) isutilized to the exclusion of directional diversity. As shown in FIG. 13,all of the antenna elements 1304 are positioned on a front facet 1308 ofthe IBR. In FIG. 13, the IBR 1300 includes eight antenna elements 1304(Q=8). It will be appreciated that the IBR 1300 may include less than ormore than eight antenna elements 1304.

FIG. 14 illustrates another embodiment of an IBR 1400 where directionaldiversity is present. IBR 1400 includes the same number of antennaelements as the IBR 1300 shown in FIG. 13 (Q=8, or 16 if usingcross-polarization feeds to all antenna elements). In FIG. 14, theantenna elements 1404 are arranged on a front facet 1408 and two sidefacets 1412. In FIG. 14, the side facets 1412 are at a 45° angle in theazimuth relative to the front facet 1408. It will be appreciated thatthis 45° angle is arbitrary and different angles are possible dependingon the specific radiation patterns of the various antenna elements.Furthermore, the angle may be adjustable so that the side facets 1412can vary in azimuth angle relative to the front facet between 0° to 90°(any value or range of values between 0° to 90°). Conventionalelectromechanical fabrication elements could also be used to make thisside facing angle dynamically adjustable by, for example, the RRC 660 ofFIG. 6 or the same in combination with the IBMS Agent 700 of FIG. 7.Additionally, variations of the embodiment of FIG. 14 can use more thanthree facets at different angular spacing all within a nominal azimuthalrange of approximately 180°, and the number of antenna elements 1404 maybe less than or greater than Q=8. For example, in one embodiment, theantenna array includes four facets uniformly distributed in an azimuthalangular range across 160°.

FIG. 15 illustrates an IBR 1500 having an “omni-directional” (in theazimuth) array of antenna elements 1504. In FIG. 15, Q=16 antennaelements 1504 are uniformly distributed across all 360° of azimuthangles, amongst the eight facets 1508-1536. Such an embodiment can beadvantageous for propagation environments with severe obstructionsbetween IBRs in a radio link or for an omni-directional common node at apoint of aggregation (i.e. fiber POP) within a PMP deployment of IBRs.It will be appreciated that the IBR may have less than or more thaneight facets, and that the number of antenna elements 1504 may be lessthan or greater than Q=16. It will also be appreciated that the antennaelements 1504 may be distributed non-uniformly across the facets.

With reference back to FIGS. 10-12, the IBR RF Switch Fabric 1012provides selectable RF connections between certain RF-Tx-m and/orRF-SW-Tx-q combinations and certain RF-Rx-n and/or RF-SW-Rx-qcombinations. In an embodiment where Q=M=N, the IBR RF Switch Fabric1012 can be parallel through connections or single-pole, single-throw(SPST) switches. In a maximally flexible embodiment where Q>Max (M, N)and any RF-Tx-m or RF-Rx-n can connect to any respective RF-SW-Tx-q orRF-SW-Rx-q, then a relatively complex cascade of individual switchblocks and a more extensive decoder logic may be required. Because eachRF-Tx-m or RF-Rx-n can be readily interchangeable amongst theirrespective sets of signals at the digital baseband level, it isgenerally only necessary to connect any given RF-Tx-m or RF-Rx-n to asubset of the Front-ends 1008 roughly by the ratio respectively of Q/Mor Q/N on average.

For example, if the IBR has Q=8 antenna elements and M=N=4, thenQ/M=Q/N=2. Thus, any of the RF-Tx-m (m=1, 2, 3, 4) signals may beconnectable to a pair of RF-SW-Tx-q signals, via a selectable RFconnection including a SPDT switch (and similarly for RF-Rx-n toRF-SW-Rx-q). In this example, either RF-Tx-m and/or RF-Rx-n couldconnect via such a selectable RF connection to either one of thefront-facing antenna elements or one of the side-facing antenna elementssuch that each RF signal has directional as well as spatial diversityoptions while allowing any two adjacent elements in the azimuthdirection to both be selected. Similarly, for the IBR shown in FIG. 15,if the IBR has Q=16 (non-polarized) antenna elements and M=N=4, then anyRF-Tx-n or RF-Rx-n signal could be oriented in one of four directions at90° increments via a selectable RF connection including a single-pole,quadrature throw (SP4T) switch.

An alternative embodiment of the IBR RF Switch Fabric 1012 can alsooptionally connect, via a signal splitter, a particular RF signal(typically one of the RF-Tx-m signals) to multiple Front-ends 1008 andantenna elements 652 simultaneously. This may be advantageous in someIBR operational modes to provide an effectively broader transmitradiation pattern either in normal operation or specifically for certainchannel estimation or broadcast signaling purposes. In context of theSPDT switch implementation in the example above for the IBR of FIG. 14,this would entail, if used for RF-Tx-m, the addition of another SPDTswitch and three passive splitter/combiners as well as decoder logic foreach antenna element pair.

In all of the foregoing descriptions of the IBR RF Switch Fabric 1012,substantially conventional components and RF board design structures asare well known can be used to physically implement such selectable RFconnections. Alternatively, these selectable RF connections can also berealized by custom integrated circuits on commercially-availablesemiconductor technologies.

With reference back to FIGS. 6 and 7, the IBR RF 632 also includestransmit and receive chains 636, 640. Exemplary transmit and receivechains 636, 640 are shown in FIGS. 16 and 17 respectively. In oneembodiment, as shown in FIG. 16, the transmit chain 636 takes a transmitchain input signal such as digital baseband quadrature signals I_(Tm)and Q_(Tm) and then converts them to a transmit RF signal RF-Tx-m.Typically, each transmit chain Tx-m 636 includes at least two signalDACs, channel filters, a vector modulator, a programmable gainamplifier, an optional upconverter, and at least one synthesized LO.Similarly, as shown in FIG. 17, the receive chain 640 converts a receiveRF signal RF-Rx-n to a receive chain output signal such as digitalbaseband quadrature signals I_(Rn) and Q_(Ra). Typically, each receivechain Rx-n 640 includes an optional downconverter, a vector demodulator,an AGC amplifier, channel filters, at least two signal ADCs and at leastone synthesized LO. A common synthesized LO can often be shared betweenpairs of Tx-m and Rx-n chains, or even amongst a plurality of suchpairs, for TDD operation IBRs. Examples of commercially availablecomponents to implement the IBR RF chains include the AD935x family fromAnalog Devices, Inc. Numerous other substantially conventionalcomponents and RF board design structures are well known as alternativesfor realizing the Tx-m and/or Rx-n chains whether for TDD or FDDoperation of the IBR.

With reference back to FIGS. 6 and 7, the specific details of the IBRModem 624 and IBR Channel MUX 628 depend somewhat on the specificmodulation format(s) deployed by the IBR. In general, the IBR requires amodulation format suitable for a broadband channel subject tofrequency-selective fading and multipath self-interference due to thedesired PHY data rates and ranges in obstructed LOS propagationenvironments. Many known modulation formats for such broadband channelsare possible for the IBR. Two such modulation formats for the IBR are(1) Orthogonal Frequency Division Multiplexing (OFDM) and (2)Single-Carrier Frequency Domain Equalization (SC-FDE). Both modulationformats are well known, share common implementation elements, and havevarious advantages and disadvantages relative to each other.

As is well known, OFDM essentially converts the frequency-selectivefading broadband channel into a parallel collection of flat-fadingsubchannels wherein the frequency spacing between subchannels is chosento maintain orthogonality of their corresponding time domain waveforms.In OFDM, a block of information symbols is transmitted in parallel on ablock of discrete frequency subchannels, each conveying one informationsymbol which can be efficiently channel multiplexed into the time domainby using an Inverse Discrete Fourier Transform (IDFT). A cyclic prefixof length in time greater than the dominant time delays associated withmulti-path self-interference is then pre-pended to the IDFT output blockby serially transmitting first in time a fraction of the IDFT outputblock time domain samples that are transmitted last. This length in timeis also sometimes called a guard interval. The use of a cyclic prefixeffectively converts a linear convolution of the transmitted block ofsymbols to a circular convolution such that the effects of inter-symbolinterference (ISI) associated with multipath time delays can be largelyeliminated at the OFDM receiver. At the OFDM receiver, the cyclic prefixis discarded and each time domain input block of symbols isdemultiplexed back into frequency domain subchannels each conveying oneinformation symbol by using a Discrete Fourier Transform (DFT). Thetransmission of a known training block of symbols within each OFDM PPDUenables the OFDM receiver to correct for carrier frequency offsets anddetermine a complex weighting coefficient for each frequency subchannelthat can equalize the effects of frequency-selective relative gain andphase distortion within the propagation channel. Furthermore,transmission of known “pilot” sequences of symbols at certainpredetermined subchannels within the transmit block enables the OFDMreceiver to track such channel distortions and frequency offsets duringreception of information symbol blocks the PPDU as well as provide acoherent reference for demodulation. Note that for those subchannelssubjected to severe flat fading, as will occur inevitably in a broadbandobstructed LOS propagation channel, the information within suchsubchannels cannot be directly demodulated. Thus to avoid a significantirreducible bit-error rate (BER) that would be unacceptable for most IBRapplications, it is essential that either forward error correction (FEC)encoding be applied with a constraint length comparable to the number ofbits per OFDM block of information symbols or with a combination ofconstraint length and interleaving depth such that related FEC encodedbits or symbols span substantially all of the OFDM block of informationsymbols.

In an SC-FDE transmitter, every block of information symbols, eachmapped to the same single carrier frequency at a relatively high symbolrate, has a cyclic prefix prepended to it prior to transmission. Similarto OFDM, the cyclic prefix consists of a finite fraction of themodulated symbols with a length in time greater than the dominant timedelays associated with multipath self-interference wherein suchmodulated symbols are identical to those to be transmitted last in timefor each block. Analogously to OFDM, this cyclic prefix effectivelyconverts a linear convolution of the transmitted block of symbols to acircular convolution such that inter-block interference (IBI) due tomultipath can be largely eliminated at the SC-FDE receiver. The SC-FDEreceiver is similar to the OFDM receiver in that a cyclic prefix removerdiscards a cyclic prefix for each block of information symbols and theremaining sampled signals are decomposed into a set of frequencysubchannels collectively representing the IBI-free block of symbolsusing a DFT. Based on a set of complex weighting coefficients, one foreach frequency sub-channel, as usually determined from a known trainingblock of symbols within each SC-FDE PPDU, the broadband channel inducedrelative distortions of amplitude and phase for each frequencysub-channel are then corrected in the Frequency Domain Equalizer (FDE).In contrast to OFDM where FDE-corrected subchannel estimates can bedirectly demapped as individual information symbols, in SC-FDE theFDE-corrected frequency domain subchannel estimates are thenre-multiplexed into a channel equalized single-carrier stream ofinformation symbol estimates using an IDFT so that such informationsymbol estimates can be subsequently demodulated.

Embodiments of the IBR may use Quadrature Amplitude Modulation (QAM) tomap groups of information data bits to a symbol including an I (or“real”) component and a Q (or “imaginary”) component. These symbols(i.e., symbols that include I and Q components) are typically referredto as “complex” symbols. Such “complex” symbols may be multiplexed ordemultiplexed by an IDFT or DFT respectively implemented by structuresexecuting a complex Inverse Fast Fourier Transform (IFFT) or complexFast Fourier Transform (FFT). In IBR embodiments, references to IDFT orDFT herein assume that such transforms will typically be implemented bystructures executing an IFFT or FFT respectively. Note also that thecyclic prefix described above can also be implemented as a cyclicpostfix for either OFDM or SC-FDE with equivalent performance. ForSC-FDE, some re-ordering of samples at the receiver after removal duringthe guard interval may be required if a cyclic postfix is used. It willbe appreciated however that techniques other than QAM for modulationmapping may also be used.

With reference again to FIGS. 6 and 7, the specific details of the IBRModem 624 and IBR Channel MUX 628 also depend somewhat on the specificantenna array signal processing format(s) deployed by the IBR. Ingeneral, the IBR utilizes multiple antennas and transmit and/or receivechains which can be utilized advantageously by several well-knownbaseband signal processing techniques that exploit multipath broadbandchannel propagation. Such techniques include Multiple-Input,Multiple-Output (MIMO), MIMO Spatial Multiplexing (MIMO-SM), beamforming(BF), maximal ratio combining (MRC), and Space Division Multiple Access(SDMA).

In general, any configuration where a transmitter that has multipletransmit antennas (or “inputs” to the propagation channel) communicateswith a receiver that has multiple receive antennas (or “outputs” fromthe propagation channel) can technically be described as MIMO. However,typically the term MIMO is used in the context of spatially multiplexingmultiple encoded and modulated information streams from multipletransmit antennas into a multipath channel for receipt at multiplereceive antennas wherein inversion of channel transfer matrix, usuallydetermined by a known training block associated with each PPDU, enablesseparation and demodulation/decoding of each information stream—aprocess called MIMO-SM. Various embodiments of the IBR, as describedherein, advantageously use other types of antenna diversity such asdirectional or polarization diversity to realize MIMO-SM performanceeven in propagation environments where spatial separation alone may beinadequate for conventional MIMO-SM.

For a given encoded and modulated information stream, BF or MRC can beutilized at either the transmitter or the receiver (or both) to improveeither the signal to interference and noise ratio (SINR) or the signalto noise ratio (SNR). For example, BF or MRC optimally combine theinformation signal received from the multiple antennas or split theinformation stream signal as transmitted by the multiple antennas.Numerous algorithms for determining the optimal weighting criteriaamongst the multiple antennas, usually as a function of frequency withina frequency-selective multipath broadband channel, are well known.

SDMA allows an Aggregation End IBR (AE-IBR) in a PMP configuration (seeFIG. 5) to transmit to or receive from multiple Remote End IBRs(RE-IBRs) simultaneously using parallel encoded and modulatedinformation streams each associated with multiple antenna andtransmit/receive chain combinations wherein stream-specific BF maximizessignal quality for a given stream and IBR pair, while minimizinginterference to other streams and IBR pairs. To the extent that multipleRE-IBRs are separable in space, or another antenna array characteristicsuch as polarization or direction, then significant increases in overallspectrum efficiency of a PMP system are possible using SDMA. As with BF(and MRC), numerous algorithms for computing the optimal weightingcriteria, such as Eigen Beam Forming (EBF), amongst multiple antennasare well known.

In view of the foregoing exemplary modulation format and antenna arrayprocessing format alternatives for the IBR, exemplary embodiments of theIBR Modem 624 and IBR Channel MUX 628 are described with reference toFIGS. 18-30.

FIG. 18 shows an exemplary embodiment of the IBR Modem 624. The IBRmodem 624 includes a Tx PLCP generator 1804, an Rx PLCP analyzer 1808,multiple modulator cores 1812 and multiple demodulator cores 1816. FIG.18 is divided functionally into a transmit path and a receive path. Thetransmit path includes the TxPLCP 1804 and the modulator cores 1812, andthe receive path includes the Rx PLCP analyzer 1808 and the demodulatorcores 1816.

The transmit path of the IBR Modem 624 includes a total of Jmod“Modulator Cores” 1812, each denoted as Modulator Core j wherein j=1, 2,. . . , Jmod. The Tx PLCP generator 1804 provides transmit datainterface streams, Tx-j, to each Modulator Core j 1812 such that Ajtotal vector outputs of mapped transmit symbol streams (denoted by the“+” on such I, Q connections) are generated from each Modulator Core j1812. This results in a total number of transmit symbol streams,K=A1+A2+ . . . +AJmod, each in vector format (I_(TS), Q_(TS))_(k) fromk=1 to K that are connected to the transmit path of the IBR Channel MUX628.

Similarly, the receive path of the IBR Modem 624 includes a total ofJdem “Demodulator Cores” 1816 each denoted as Demodulator Core j 1816wherein j=1, 2, . . . , Jdem. The IBR Channel MUX receive path providesL=B1+B2+ . . . +BJdem vector format receive symbol streams (I_(RS),Q_(RS))_(l) for l=1 to L that are input as Bj vector streams per eachDemodulator Core j 1816 to produce the receive data interface streamRx-j.

In PTP IBR configurations where Jmod=Jdem, usually Aj=Bj. However, for aPTP IBR where probing capability is present, Aj=Bj only for j=1 to Jmodin cases where Jdem>Jmod. In PMP IBR configurations, it may happen thatAj≠Bj; even if Jmod=Jdem.

An exemplary embodiment of a Modulator Core j 1812 is illustrated inFIG. 19. As shown in FIG. 19, this exemplary modulator core 1812includes a scrambler 1908, encoder 1912, stream parser 1916, multipleoptional interleavers 1920, multiple symbol groupers 1924 and multiplesymbol mappers 1928.

Typically, the data from the Tx PLCP generator 1804, Tx-j 1904, isscrambled at the scrambler 1908 and then passed to the encoder 1912. Theencoder 1912 provides FEC encoding and in some types of encoders alsoeffectively interleaves the encoded data. Exemplary FEC encoder typesfor IBRs include convolutional, turbo and low density parity check(LDPC). The encoded data is passed to the Stream Parser 1916. The StreamParser 1916 demultiplexes the encoded data into Aj streams. Each encodeddata stream is then interleaved if necessary at the optionalinterleavers 1920 such that the greater of the FEC encoder constraintlength and the interleaving depth approximates the total number of bitsper transmitted block of symbols in either OFDM or SC-FDE. Exampleinterleaver types include block (or “row/column”) and convolutional.Such interleaved and/or encoded data in each stream is then grouped atsymbol groupers 1924 based on the number of encoded bits per symbol. Thegroups of bits pass to the symbol mapper 1928, which converts each groupof encoded bits into a vector representation (I, Q) within aconstellation and then provides an output as a transmit symbol stream1932. An exemplary technique for mapping encoded bits to a vector symbolis QAM optionally with Gray coding.

FIG. 20 illustrates an exemplary embodiment of a Demodulator Core j 1816compatible with the exemplary Modulatory Core j 1812 of FIG. 19. Asshown in FIG. 20, the demodulator core 1816 includes a descrambler 2008,decoder 2012, stream MUX 2016, multiple optional deinterleavers 2020,and multiple soft decision symbol demappers 2024.

The Demodulator Core j 1816 at the highest level can be described asperforming essentially the reverse operations of those performed in theModulator Core j 1812 of FIG. 19. A key difference though is that thevector representation (I, Q) receive symbol streams 2032 input to eachDemodulator Core j 1816 are only estimates which may be corrupted due tochannel or receiver impairments such as multipath self-interference,Gaussian noise, co-channel interference, carrier frequency offset (CFO),distortion through channel filters at Tx and/or Rx, non-linearity in theTx and/or Rx chains and Front-ends, or phase noise in either Tx and/orRx local oscillators. Thus, the Demodulator Core j 1816 may use the softdecision symbol demapper 2024, which estimates the likelihood that areceived symbol or bit has a particular value using, for example, aknown technique such as Log-Likelihood Ratio (LLR). For example, if eachdata bit out of the demapper 2024 had a soft-decision representation (or“metric”) of 8 bits, then a value of 0 would represent a data bit of 0,a value of 1-20 would indicate most likely a data bit of 0, a value of21-125 would indicate more likely a data bit of 0 than a data bit of 1,a value of 126-129 would indicate near uncertainty of the data bit aseither 0 or 1, a value of 130-234 would indicate more likely a data bitof 1, a value of 235-254 would indicate most likely a data bit of 1, anda value of 255 would represent a data bit of 1. These soft-decisionmetrics can then be deinterleaved if applicable at optionaldeinterleavers 2020 and stream multiplexed at the Stream MUX 2016, andthen supplied to the decoder (deinterleaver) 2012 as a sequence ofsoft-decision metrics estimating the originally encoded (and possiblyinterleaved) bit stream. Decoder types are matched to encoder types asis well known. Techniques such as iterative decoding within the IBRmodem or combination of IBR modem and IBR Channel MUX are also known andcan be used in some embodiments.

FIGS. 19 and 20 are examples of implementations of the modulator anddemodulator cores 1812, 1816, respectively, in the IBR Modem 624. Itwill be appreciated that the order of the Scrambler 1908 and Encoder1912 can be reversed (and hence the Decoder 2012 and Descrambler 2008could be reversed). Also, the Stream Parser 1916 can divide a transmitdata interface stream Tx-j sequentially into multiple encoder/scramblercombinations and hence the corresponding Stream Mux 2016 would combinemultiple Decoder outputs. FIGS. 21 and 22 illustrate exemplaryalternatives with the above recited differences and without showingoptional separate interleavers 1920 and deinterleavers 2020.

Although each element is illustrated separately in FIGS. 19-22, theelements are not required to be distinct physical structures. It isfeasible to time multiplex either the entire Modulator or DemodulatorCores 1816, 1820 or constituent elements such as the Encoder 1912 orDecoder 2012 across multiple streams. Furthermore, in TDD operation, itcan be advantageous to time multiplex constituent elements such as theEncoder 1912 and/or Decoder 2012 even during times where the oppositetransmit/receive path is over the air compared to the time multiplexedelement. For example, by buffering receive symbol stream samples fromthe receive path of the IBR Channel MUX 628 during the receive portionof TDD operation, some of the streams can be decoded in a common Decoder2012 even when the IBR has changed over to transmit operation.

Note further that the exemplary embodiments of FIGS. 19-22 for theModulator and Demodulator Cores of the IBR Modem are all suitable foreither OFDM or SC-FDE operation.

FIG. 23 (FIGS. 23A and 23B together) illustrates an exemplary embodimentof the IBR Channel MUX 628. FIG. 23 is divided functionally into atransmit path (FIG. 23A) and a receive path (FIG. 23B). In FIG. 23A, thetransmit path includes multiple block assemblers 2304, multiple transmitchannel equalizers (CEs) 2308, multiple transmit MUXs 2312, multiplecyclic prefix addition (CPA) and block serializers 2316, multipletransmit digital front ends (DFEs) 2320, an optional preamble sampleslibrary 2324, and a pilot (and optionally preamble) symbol library 2328.In FIG. 23B, the receive path includes an acquisition/synchronizationcalculator 2336, multiple receive digital front ends (DFEs) 2340,multiple cyclic prefix removal (CPR) and block assemblers 2344, multiplecomplex DFTs 2348, multiple receive channel equalizers 2352 and multiplereceive MUXs 2356. The IBR Channel MUX 628 also includes a channelequalizer coefficients generator 2332 (see FIG. 23A). The channelequalizer coefficients generator 2332 may be used by both transmit andreceive paths or only the receive path depending on the specificembodiment. In the receive path, the IBR Channel MUX 628 is frequencyselective to enable operation in obstructed LOS propagation environmentswith frequency selective fading within the channel bandwidth as well asoperation in predominantly unobstructed LOS.

In the transmit path, each transmit symbol stream of mapped symbols(I_(TS), Q_(TS))_(k) from k=1 to K 2360 is connected to a respective TxBlock Assembler k 2304 as shown in FIG. 23A. For OFDM modulation, eachTx block Assembler k 2304 is equivalent to a serial to parallel bufferthat also places “running” pilot symbols in pre-determined locationscorresponding to pilot subchannels. In the exemplary embodiment of FIG.23A, such pilot symbols are supplied to each Tx Block Assembler k 2304by a Pilot Symbol Library 2328. Alternatively, pilot symbols areselectively injected into the data stream at predetermined points with aModulator Core 1812 similar to FIG. 19 or 21 but with a “Pilot DataLibrary” (not shown) providing grouped data to a selectable buffer ormultiplexer (not shown) between the Symbol Grouper 1924 and SymbolMapper 1928 of either FIG. 19 or 21.

For SC-FDE modulation with no frequency selective channel equalization,the Tx Block Assembler k 2304 would, as an exemplary embodiment, be asimple serial to parallel buffer that places pilot symbols inpre-determined symbol sequence positions either from the Pilot SymbolLibrary 2328 as shown, or alternatively from a “Pilot Data Library” inthe Modulator Core as described above. If frequency selective channelequalization in Tx for SC-FDE is used, then in addition to the above,the Tx Block Assembler k 2304 would also include a Complex DFT thatfollows the serial to parallel buffer. In this DFT structure version ofSC-FDE (also known as DFT pre-coding SC-DFE), it is also possible toinsert the pilot symbols as “pseudo-subchannels” after the DFT operationinstead of as time domain symbols before the DFT.

In alternative embodiments that utilize frequency selective channelequalization in Tx for SC-FDE, the transmit channel equalizer 2308 mayutilize a time domain approach as shown in FIG. 24A. The embodimentshown in FIG. 24A does not require the Complex DFT as part of the TxBlock Assembler k 2304 shown in FIG. 23A because the transmitequalization is performed in the time domain, obviating the need for theDFT or the inverse DFT (IFFT).

After block assembly, the blocks of mapped symbols (each a vector of (I,Q) constellation points) are typically supplied to each of M transmitchannel equalizers 2308 (“Tx-CE-m” for m=1 to M in FIG. 23A). EachTx-CE-m 2308 applies K blocks of amplitude and/or phase weights (eachamplitude phase weight may be represented in Cartesian form as a“complex” weight with a “real” and “imaginary” sub-component values)respectively to each of the K blocks of mapped vector symbols.

An exemplary embodiment of a transmit channel equalizer Tx-CE-m 2308 isshown in FIG. 24. The blocks of mapped vector symbols are {right arrowover (TxBlock)}_(k) wherein each symbol element within the vector {rightarrow over (TxBlock)}_(k) has an (I, Q) or, relative to FIG. 24,“complex” value (I representing a “real” sub-component value, Qrepresenting an “imaginary” sub-component value). Each {right arrow over(TxBlock)}_(k) vector of complex symbols is then complex multiplied by arespective transmit weight vector {right arrow over (WT)}_(m,k) as shownin FIG. 24. The channel equalized vector of symbols for each transmitchain m is then {right arrow over (TxBlockEq)}_(m) (the transmit chainchannel-equalized symbols) which represents the complex summation of thesymbol by symbol output vectors of each respective weighting operation.Note that FIG. 24 depicts an exemplary implementation of transmitchannel equalizer TX-CE-m 2308 based on vector complex multiplicationand summation. Such an approach is easily implemented using eithergeneric function calls or custom software modules executing on anembedded processor (or Digital Signal Processor—“DSP”). However,numerous other implementation techniques are also known such asdedicated logic circuits for complex multiplication and/or summation, aset of four scalar multiplier circuits and two scalar adders, or otherlogic circuits such as combinatorial gates (i.e. OR, NOR, AND, NAND,XOR, XNOR, etc.), multiplexers, shift registers, etc. that can producean equivalent result.

An alternative exemplary embodiment of a transmit channel equalizerTx-CE-m 2308 is shown in FIG. 24A. The transmit channel equalizerTx-CE-m 2308 allows for time domain based processing. The transmitchannel equalizer Tx-CE-m 2308 is similar to the TX-CE-m shown in FIG.24, but not need a DFT in the Tx Block Assembler k 2304 or an IDFTwithin the Tx-Mux-m 2312. As described in relation to FIG. 24, theblocks of mapped vector symbols are {right arrow over (TxBlock)}_(k)wherein each symbol element within the vector {right arrow over(TxBlock)}_(k) has an (I, Q) or, relative to FIG. 24, “complex” value (Irepresenting a “real” sub-component value, Q representing an “imaginary”sub-component value). In the embodiment of FIG. 24A, the processing ofthese complex values is performed differently, having a distinctadvantage in complexity for a lower number of channel equalizationimpulse response length relative to the traditional approach forfrequency domain equalization as described in relation to FIG. 24. Each{right arrow over (TxBlock)}_(k) vector of complex symbols passedthrough a complex finite impulse response filter (FIR) 2401. Suchstructures and DSP processing implementations are well known in the artand comprise a mathematical convolution with the FIR filter weights.Such weights result in a desired impulse response and are provided tothe FIR 2401 by the transmit weight vector {right arrow over (WT)}_(m,k)as shown in FIG. 24A. A circular summer 2405 is also used to convert thelinear convolution of 2401 to a circular convolution. The circularconvolution retains the original number of K symbols of the {right arrowover (TxBlockEq)}_(m), rather than expanding them by one less than thelength of the FIR filter weights (as is the case with linearconvolution). The channel equalized vector of symbols for each transmitchain m is then {right arrow over (TxBlockEq)}_(m) (the transmit chainchannel-equalized symbols) which represents the complex summation of thesymbol by symbol output vectors of each respective circular convolutionoperation between each {right arrow over (TxBlock)}_(k) and the transmitweight vector {right arrow over (WT)}_(m,k).

The circular FIR filtering is described in detail with reference toFIGS. 24B-24I. FIG. 24B is a plot of an exemplary transmit equalizerimpulse response for use with the Tx-CE-m having a complex circular FIRcapability. FIG. 24C is a plot of an exemplary transmit equalizerfrequency response for use with the Tx-CE-m having the complex circularFIR capability.

FIGS. 24D, E, and F are presented together on the same drawing sheet forpurposes of discussion. FIG. 24D is equivalent to FIG. 24B. FIG. 24E isa plot of an exemplary 16 QAM—10 symbol Transmit Block ({right arrowover (TxBlock)}_(k)), where K=10 (K is the number symbols containedwithin an example transmit block). It will be appreciated that K istypically much larger, but a shorter block size of K=10 is utilized toeasily illustrate the processing with the Tx-CE-m having the complexcircular FIR capability. FIG. 24F shows the result of a linearconvolution of the exemplary 16 QAM—10 symbol Transmit Block ({rightarrow over (TxBlock)}_(k)) with the example complex impulse response. Alinear convolution between K samples and WT samples results in K+WT=1value. However, an expansion in the number of transmit symbols isundesirable and does not provide for the equivalent processing of afrequency domain implementation of FDE processing. Taking the WT-1 lastsamples of the resulting linear convolution output, and summing themrespectively with the first WT-1 samples of the linear convolutionoutput results in an equivalent circular convolution. Such processingmay be performed in the circular summer 2405 shown in FIG. 24A.

FIG. 24G is a plot of the result of the FIR based circular convolutionutilizing the linear FIR 2401 and the circular summer (2405) of theexemplary 16 QAM—10 symbol Transmit Block ({right arrow over(TxBlock)}_(k)) with the example complex impulse response. FIG. 24H is aplot of the result of a frequency domain based circular convolution ofthe exemplary 16 QAM—10 symbol Transmit Block ({right arrow over(TxBlock)}_(k)) with the example complex impulse response. FIG. 24G andFIG. 24H show the resulting circular convolutional processing output asperformed respectively by the circular FIR of FIG. 24A, and traditionalfrequency domain circular convolutional based processing as described inrelation to the frequency selective transmit channel equalization, whenapplied to SC-FDE. FIG. 24I is a magnitude plot of the result of boththe circular FIR and the frequency domain based circular convolutions ofthe exemplary 16 QAM—10 symbol Transmit Block ({right arrow over(TxBlock)}_(k)) with the example complex impulse response forcomparison, showing that these values are identical. Such use of thetime domain processing for frequency selective transmit equalizationeliminates the requirement for both a DFT and an IDFT, resulting inreduced processing requirements for transmit weight vectors of moderatelength.

Note further that for embodiments with either OFDM or SC-FDE modulationand no frequency selective channel equalization, {right arrow over(WT)}_(m,k) would typically be composed of a block of identical transmitweights applied equally to all mapped symbols for a given stream k. Insome OFDM or SC-FDE embodiments where K=M, no equalization or weightingamongst transmit streams and chains may be desired such that eachTx-CE-m 2308 in FIG. 23A (or FIG. 24) is equivalent to a throughconnection mapping each {right arrow over (TxBlock)}_(k) directly toeach {right arrow over (TxBlockEq)}_(m) for each k=m.

For each transmit chain Tx-m, the channel equalized vector of symbolsfor each transmit chain {right arrow over (TxBlockEq)}_(m) is suppliedto Tx-Mux-m 2312, where m=1 to M, as shown in FIG. 23A. In the case ofOFDM modulation or SC-FDE modulation with frequency-selective channelequalization utilizing the TX-EQ-m 2308 of FIG. 24, a DFT is requiredwithin the TX Block Assembler k 2304, and each TX-Mux-m 2312 includes anIDFT to transform the successive frequency domain symbols into a blockof time domain samples. Note that in the case of embodiments of theTX-EQ-m of FIG. 23A (the time domain frequency selective equalization)no such IDFT is required. For SC-FDE modulation with no frequencyselective channel equalization, and hence no DFT pre-coding within eachTx Block Assembler k 2304, then {right arrow over (TxBlockEq)}_(m) isalready effectively a block of transmit chain time domain samples and nosuch IDFT is required.

With reference back to FIG. 23A, each respective Tx-Mux-m 2312 isfollowed by a CPA and Block Serializer-m 2316. “CPA” refers to “CyclicPrefix Addition”-cyclic prefix was described above—which is performed bythe cyclic prefix adder within each CPA and Block Serializer-m 2316.Each CPA and Block Serializer-m 2316, via a cyclic prefix adder,typically replicates a set of samples from the end of the block andprepends these samples to the beginning of a cyclically-extended blockor includes logic to read out these samples in a way that produces anequivalent result. The CPA and Block Serializer-m 2316, via a blockserializer, then effectively performs a parallel to serial conversion ofthe extended block into a sequence of cyclically-extended (I, Q)transmit chain time domain samples.

The cyclically-extended block of (I, Q) time domain samples for eachtransmit chain Tx-m are then supplied to a respective Tx-DFE-m 2320 (DFErefers to Digital Front End). Each Tx-DFE-m 2320 performs a variety ofpossible time domain signal processing operations that may be specificto particular IBR embodiments. Common DFE processes in the transmit pathinclude digital up-conversion of the sampling rate, DC offsetcalibration for each Tx-m transmit chain, digital pre-distortion toaccount for non-linearities in the analog transmit path, pulse-shapingfiltering, crest factor or peak to average power ratio (PAPR) reduction,or frequency shifting to an “intermediate Frequency” (IF) suitable foranalog up conversion to RF-Tx-m in the Tx-m transmit chain (as opposedto the baseband (I_(T), Q_(T))_(m) interface illustrated in FIG. 16).Except for the last listed option of digital IF up conversion, theoutput (a transmit chain input signal) of each respective Tx-DFE-m 2320is typically a sequence of cyclically-extended blocks of calibrated,compensated and filtered baseband symbol samples (I_(T), Q_(T))_(m).

In the receive path of the exemplary IBR Channel MUX 628, downconvertedand amplified samples from I and Q ADCs in each Rx-n receive chain (areceive chain output signal) are passed to respective Rx-DFE-n 2340.Although the receive path of the IBR generally follows the logic ofreversing the operations of the transmit path, the details areconsiderably different in the IBR Channel MUX because in the receivepath the samples are corrupted by channel propagation impairments andarriving at initially unknown times. In view of this the receive pathDigital Front Ends of the IBR Channel MUX shown in FIG. 23B, Rx-DFE-n2340, perform several time domain digital operations typically includingmatched filtering, rotating, sampling rate downconversion, and Rx-nchain calibration. At minimum, the Rx-DFE-n 2340 is used in thedetection of timing synchronization information in the optional Preambleof the L PPDU-l transmissions from the transmitting IBR.

FIG. 26 illustrates this Preamble in the time domain. Note that theoptional Preamble for the l-th Tx-path in a transmitting IBR could beeither generated from a symbol library (as indicated optionally in Pilot(And Preamble) Symbol Library 2328 of FIG. 23) or read directly from aPreamble Samples Library 2324 (also indicated optionally in FIG. 23).

Referring again to FIG. 23B, the Acquisition/Synchronization detector2336 acquires a timing reference, for example, from detection of one ormore of the up to L optional preambles, one or more of the blocks ofTraining Pilots, an optional known pattern within a PLCP header, and/orpilot symbols within blocks of transmit symbol streams. TheAcquisition/Synchronization detector 2336 provides a feedback signal toeach Rx-DFE-n 2340 that enables, for example, symbol rotation andalignment in the time domain (typically in the RX-DFE-n using a CORDICalgorithm). The Acquisition/Synchronization detector 2336 furtherprovides symbol boundary and block boundary timing references (notshown) for use by various elements of the IBR Channel MUX 628 and theIBR Modem 624.

Such corrected receive symbol samples are then supplied to a respectiveCPR and Block Assembler-n 2344. “CPR” means “cyclic prefix removal.” TheCPR and Block Assembler-n 2344 effectively discards the number ofreceived samples at beginning of each block corresponding to the numberof cyclic prefix samples prepended to each block in the transmit path ofthe transmitting IBR. The remaining samples are serial to parallelbuffered into a single block (or an equivalent operation) suitable fordecomposition into receive chain frequency domain subchannels in eachrespective Complex DFT-n 2348.

As shown in FIG. 23B, the sets of receive chain frequency domainsubchannel samples output from the N Complex DFT-n 2348 are collectivelysupplied to L frequency domain receive channel equalizers 2352, each aRx-CE-l. FIG. 25 illustrates an exemplary embodiment of Rx-CE-l 2352.The receive channel equalizer Rx-CE-l 2352 is analogous to the transmitchannel equalizer Tx-CE-m 2308 shown in FIG. 24. Although implementationdetails of each Rx-CE-l 2352 can vary considerably just as describedabove for TX-CE-m 2308, essentially the frequency domain receive vectorsof I and Q samples for each block, {right arrow over (RxBlock)}_(n), arerespectively complex multiplied by complex receive weights {right arrowover (WR)}_(l,n) and then complex summed to produce a set ofchannel-equalized frequency domain estimates, {right arrow over(RxBlockEq)}_(l), representative of the original transmitted blockcorresponding to stream l.

The task of producing such complex receive weight vectors {right arrowover (WR)}_(l,n) that allow each Rx stream l to be separated from themyriad of signals, both desired and undesired, received via N (N≧L)receive chains is performed by the Channel Equalizer CoefficientsGenerator 2332 of FIG. 23A. There are many known algorithms forgenerating the appropriate frequency selective complex receive weightvectors for the IBR Channel MUX as shown in FIG. 23 wherein a known“Training Block” of pilot symbols for each stream l is used to determinethe receive weight vectors {right arrow over (WR)}_(l,n) that allowlinear detection of each {right arrow over (RxBlockEq)}_(l) vector inview of the actual receive chain frequency domain subchannel samplesobserved for the N receive chains during a reception periodcorresponding to transmission of the Training Block (or block ofTraining Pilots). In general, a Channel Equalizer Coefficients Generator2332 that is based on these algorithms calculates the various weights bycomparing the actual receive chain frequency domain subchannel samplesobserved during the reception of the Training Block with the expectedfrequency domain subchannel samples corresponding to a particular knownTraining Block for any given transmit stream. Such algorithms includeZero Forcing (ZF), Maximal Ratio Combining (MRC) and Minimum Mean SquareError (MMSE). Alternatively, other known algorithms for non-lineardetection of each {right arrow over (RxBlockEq)}_(l) are also known,such as Successive Interference Cancellation (SIC) in combination withZF, MRC or MMSE, V-BLAST (an acronym for “vertical Bell Labs LayeredSpace-Time) with ZF or MMSE, or Maximal Likelihood (ML).

The Channel Equalizer Coefficients Generator 2332 also supplies thecomplex transmit weight vectors {right arrow over (WT)}_(m,k) used fortransmit channel equalization within the exemplary IBR Channel MUX 628.In an ideal PTP TDD configuration where K=L, M=N, and no otherco-channel interference beyond the multiple transmit streams, forfrequency domain transmit channel equalization, a straightforwardalternative is to derive {right arrow over (WT)}_(m,k) directly from thecomputed {right arrow over (WR)}_(l,n) of the previous superframe.However, this can be sub-optimal for situations in which a co-channelinterferer, such as for example either another IBR PTP or PMP link inthe same vicinity or a conventional PTP link nearby, affects thereceived signals at the N receive chains of the AE-IBR differently fromthose at the RE-IBR. An alternative is to calculate {right arrow over(WR)}_(l,n) using both MMSE (which will maximize SINR at the receiver)and MRC (which will maximize SNR, or effectively maximize signal power,at the receiver) and then derive {right arrow over (WT )}_(m,k) from thecomputed, but unused in receive channel equalization, {right arrow over(WR)}_(l,n) for MRC. Note that for SC-FDE with no frequency selectivechannel equalization, then the constant {right arrow over (WT )}_(m,k)values applied to all symbols in a {right arrow over (TxBlock)}_(k) canbe blended using known algorithms from vector {right arrow over (WT)}_(m,k) values derived for MRC. This allows such an SC-FDE PTPtransmitter to improve the overall signal quality at the other receiverwhile allowing the other receiver to equalize both interference andfrequency selective fading effects. In view of the foregoing, it isadvantageous for the PMP AE-IBR to use either OFDM or SC-FDE withtransmit frequency selective channel equalization such that the ChannelEqualizer Coefficients Generator can compute MIMO-SDMA complex weights,using for example known EBF or Space Time Adaptive Processing (STAP)algorithms, to minimize multi-stream interference to RE-IBRs thatbeneficially receive only a subset of the transmitted streams. If SC-FDEwithout frequency selective transmit equalization is used, there isstill a benefit in using scalar {right arrow over (WT )}_(m,k) transmitweights at the AE-IBR derived for streams to a given RE-IBR fromprevious superframe MRC at the AE-IBR receiver but some additionalsignal separation such as time and/or frequency may be required todirect data to specific RE-IBRs.

With reference back to FIG. 23B, each respective channel equalized andinterference cancelled symbol samples stream vector {right arrow over(RxBlockEq)}_(l) (or set of channel-equalized frequency domainestimates) is then supplied to an Rx-MUX-l 2356. For OFDM, this istypically a parallel to serial converter that removes the pilotsubchannel symbol samples which are not needed in the IBR Modem. ForSC-FDE, each Rx-MUX-l 2356 includes a Complex IDFT and a parallel toserial converter that removes the time domain symbol samples at thepilot symbol sequence positions. Note though that for IBR embodimentswhere the transmitter uses SC-FDE with frequency selective equalizationand inserts the pilot symbols as “pseudo-subchannels” (see above), eachRx-MUX-l also needs to discard the pilot subchannels prior to theComplex IDFT transformation back to a time domain symbol samples stream.The output of each Rx-MUX-l is a receive symbol stream l that isprovided to a respective input to a demodulator core j.

FIG. 26 illustrates block by block generation of a given transmit symbolstream l in the transmitter IBR as combined into a transmit chain m. Thedesignator “l” is used in FIG. 26 instead of “k” for the transmitter todenote the generation of what becomes the “l-th” receive symbol streambeing demodulated at the receiver. Per the above description of FIG. 23,optionally every PPDU-l can start with a Preamble of known samples thatenables rapid signal acquisition and coarse synchronization. However,for some embodiments of the IBR in either FDD or in TDD with fixed andshort superframe timing, it is not necessary to perform such acquisitionand synchronization from the start of every PPDU. Thus, overhead can beadvantageously minimized by either discarding the Preamble from thePPDU-l generation process, at least for subsequent PPDUs after startup,or effectively combining the desired characteristics of the Preambleinto the Training Block via the Pilots-l. This approach can also be usedwith a Channel Sense Multiple Access (CSMA) approach to MAC schedulingto the extent that a timing master in the system, usually the AE-IBR,maintains sufficiently frequent timing synchronization transmissions.

In some embodiments, the PLCP Header for stream-l is matched to aparticular block such that the modulation and coding scheme (MCS) ofsuch PLCP Header block (see, for example, Block 1 of FIG. 26) is alwayspredetermined and usually chosen to deliver the PLCP header with higherreliability than the data payload blocks to follow. In typicalembodiments, the MCS includes an index number that conveys informationregarding parameters used in the modulation mapping and the forwarderror correction encoding processes. In such embodiments, the PLCPHeader typically conveys at least the information needed to inform thereceiver at the receiving-end IBR of the MCS used for the data payloadblocks to follow. In other embodiments, the MCS or informationindicating an increment, decrement or no change amongst a list ofpossible MCS values is conveyed within Block 0 of FIG. 26 by, forexample, adjusting the modulation of certain pilot symbols (or pilotsubchannels). In the case where Block 1 of FIG. 26 can carry more datathan just the PLCP Header alone, then the IBRs may allocate PPDU payloadto Block 1 even though the bits may be transferred at a different MCSthan that of subsequent Blocks.

In some embodiments, each Block 1 through f (or Block 1 through r) maycorrespond to the output of a block encoder and/or block interleaver ofFIG. 19 or 21. “f” denotes a forward link transmitted from an AE-IBR toan RE-IBR and “r” denotes a reverse link transmitted from an RE-IBR toan AE-IBR. Furthermore, each Block may include error detection bits(such as an FCS) or error correction bits (such as for a Reed-Solomon ora Reed-Muller outer encoder) that can be used either for an AutomaticRepeat reQuest (ARQ) re-transmission protocol of certain blocks or forinput to an RLC and/or RRC entity at either the AE-IBR or RE-IBR tooptimize future link parameters or resource selections.

FIG. 27 illustrates an exemplary embodiment of the Tx PLCP generator1804 of the IBR modem of FIG. 18. As shown in FIG. 27, the Tx PLCPgenerator 1804 includes a Tx PLCP Demux 2704 and multiple Tx PLCPModulators 2708. Based on input from the RRC 660, the Tx PLCP Demux 2704provides data interface streams TxD-j 2716, for j=1 to Jmod, torespective TxPLCP Mod-j 2708 that provide transmit data interfacestreams Tx-j 2720 to a respective Modulator Core j 1812 (shown in FIG.18).

In a PTP IBR configuration, such a plurality of data interface streams2716 and Modulator Cores 1812 may be used, for example, to provide linkdiversity, such as different sets of RLC parameters, carrier frequenciesand/or antenna selections where each set has appropriately multiplexedstreams and chains, or to provide probing capability. In a PMP IBRconfiguration, in addition to the above for PTP, such a plurality ofdata interface streams and Modulator Cores may also be used at an AE-IBRto optimize data transfer to certain RE-IBRs (or groups of RE-IBRs) byusing techniques such as SDMA.

FIG. 28 illustrates an exemplary embodiment of the TxPLCP Mod-j 2708 ofFIG. 27. As shown in FIG. 28, the TxPLCP Mod-j 2708 includes a PLCPheader generator 2804, a PLCP input buffer 2808, a PLCP controller 2812,a training data library 2816, a padding generator 2820 and a PLCPtransmit MUX 2824. In FIG. 28, the PLCP Controller 2812 takes input fromthe RLC and the RRC as well as the current state of the PLCP InputBuffer 2808 to perform “rate matching” for a given PPDU. Rate matchingtypically is used to determine a particular choice of modulation andcoding scheme (MCS) in view of the instantaneous data transfer demandand recent link conditions. In addition to choosing an MCS, the PLCPController 2812 supplies information to the PLCP Header Generator 2804and directs the PLCP Input Buffer 2808 and the Padding Generator 2820 tosupply data to the PLCP Transmit Mux 2824. Depending on the status ofthe RRC and the instantaneous data transfer demand, the PLCP Controller2812 also enables training data from the Training Data Library 2816 tobe multiplexed into a given PPDU as indicated by certain fields in thePLCP Header. Typically, in some embodiments, the PLCP Header conveys atleast the MCS of the PPDU after Block 1, the length of the PPDU payload(for example, either in bytes of payload or in number of blocks pluslength in bytes of padding), the MCS of any training data, the selectedtraining dataset, and the number of training data blocks. Optionally,reply acknowledgements (ACKs) or non-acknowledgements (NACKs) for one ormore previous receive PPDUs corresponding to a companion receive datainterface stream j may also be sent in the PLCP Header. To the extentthat the PLCP also concatenates multiple MPDUs and/or fragments thereof,the PLCP Header Generator also includes sufficient information to allowthe RxPLCP to recover such MPDUs and/or fragments upon receipt. Asdescribed above, FIG. 26 illustrates an exemplary PPDU having a PLCPHeader, a PPDU payload including one or more MPDUs or fragments thereof,and a PAD to cause the PPDU to occupy an integer number of Blocks asshown. Not shown in FIG. 26 is a PPDU extension with one or more Blocksof data from the Training Data Library 2816 of FIG. 28 as may beindicated by the PLCP Header.

Note that for Modulator Cores 1812 that divide each transmit datainterface stream Tx-j into two or more transmit symbol streams, the PLCPHeader of FIG. 28 may be an amalgam of two or more PLCP Headers forrespective transmit symbol streams especially in the case where aseparate MCS applies to each stream (the “vertical encoding” case shownfor example in FIG. 21). Alternatively, a single PCLP Header may be usedper Modulator Core j 1812 with parts of the PLCP Header apportionedamongst two or more transmit symbol streams (the “horizontal encoding”case shown for example in FIG. 19). Another alternative would have asingle PLCP Header encoded and replicated across all transmit symbolstreams of a particular Modulator Core j 1812. In some embodiments, eachPLCP Header further includes an FCS (not shown in FIG. 28) to enable theTx PLCP to determine if the PLCP Header has been demodulated and decodedwithout error. Note also that the PLCP Controller 2812 may be used incertain embodiments to provide or control the provision of timingsignals to the IBR Channel MUX 628 (not shown in FIG. 23) that enableinsertion of either or both of the optional Preamble or of the block ofTraining Pilots as shown in FIG. 26. In exemplary embodiments, a blockof Training Pilots, used, for example, to update transmit and/or receiveweights by the IBR at an opposite end of a link, may be inserted at thebeginning of every transmit superframe, the beginning of every PPDU (asshown in FIG. 26), or at such intervals within a PPDU or superframe asmay be communicated to the Tx PLCP generator 1804 or IBR Channel MUX 628via the RRC 660.

FIG. 29 illustrates an exemplary embodiment of the Rx PLCP analyzer 1808of FIG. 18. As shown in FIG. 29, the Rx PLCP analyzer 1808 includes a RxPLCP MUX 2904 and multiple Rx PLCP Demod-j 2908. In FIG. 29, multiplereceive data interface streams Rx-j each from respective DemodulatorCore j 1812 (see FIG. 18) are input to a respective Rx PLCP Demod-j2908. After removing the respective PLCP Headers, Padding (if any), andTraining Data (if any), each RxD-j is then multiplexed in the RX PLCPMux 2904 and transferred to the IBR MAC 612 as Rx Data 620.

FIG. 30 illustrates an exemplary embodiment of the Rx PLCP Demod-j 2908of FIG. 29. As shown in FIG. 30, the RxPLCP Demod-j 2908 includes a PLCPheader analyzer 3004, a PLCP output buffer 3008, a PLCP controller 3012,a training data analyzer 3016, and a PLCP receive Demux 3024. The PLCPController 3012 may be shared in some embodiments with the PLCPController 2812 of FIG. 28. The Tx or Rx PLCP Controllers 2812, 3012 mayalso be shared across some or all data interface streams. When a PPDU isreceived from a receive data interface stream Rx-j, the PLCP HeaderAnalyzer 3004 filters certain fields from the PLCP Header and sends thisinformation to the PLCP Controller 3012 (or alternatively, is a part ofthe PLCP Controller 3012). This enables the exemplary PLCP Controller3012 to signal the MCS for subsequent symbols or blocks to DemodulatorCore j 1812 as shown in FIG. 30 and to control the PLCP Receive Demux3024 such that the receive data interface stream Rx-j is directed to thePLCP Output Buffer 3008 or Training Data Analyzer 3016 or ignored in thecase of Padding. Further information from the PLCP Header Analyzer 3004and/or PLCP Controller 3012 enables the Training Data Analyzer 3016 todetermine the match between transmitted and received/demodulated/decodedtraining data and communicate certain metrics such as the number of biterrors and the distribution of errors to the RRC 660. In embodimentswhere the PLCP header carries an FCS, the PLCP Header Analyzer 3004 alsochecks the FCS (not shown in FIG. 30) and if a failure occurs, signalsat least the PLCP Controller 3012, the RRC 660, and the RLC 656. ThePLCP Controller 3012 may optionally also inhibit sending RxD-j for theinstant PPDU in the case of such an FCS failure or it may send it onwith an indicator (not shown) to the IBR MAC that such an FCS failurehas occurred.

With reference back to FIGS. 6 and 7, other IBR elements include the IBRMAC 612, the Radio Link Control (RLC) 656, the Radio Resource Control(RRC) 660 and, specific to FIG. 7, the IBMS Agent 700. Although IBRembodiments are possible wherein the MAC 612, RLC 656, RRC 660 and IBMS700 are distinct structural entities, more commonly IBRs are realizedwherein the IBR MAC 612, RLC 656, RRC 660, IBMS 700 and portions of theIBR Interface Bridge 608 are software modules executing on one or moremicroprocessors. Note also that in some IBR embodiments that use of a“Software Defined Radio” (SDR) as the IBR Modem 624 and/or IBR ChannelMUX 628 or portions thereof may also be realized in software executingon one or more microprocessors. Typically in SDR embodiments, the one ormore microprocessors used for elements of the PHY layer are physicallyseparate from those used for the MAC 612 or other layers and arephysically connected or connectable to certain hardware cores such asFFTs, Viterbi decoders, DFEs, etc.

FIGS. 31 and 32 illustrate exemplary views of the IBRs of FIGS. 6 and 7,respectively, showing exemplary communications protocols stacks. It willbe appreciated that some embodiments have the one or more 802.1 MACinstances, the IBR LLC, the IBR MAC, the RRC, the RLC, the IBMS and allother upper layer protocols implemented as software modules or processesexecuting on one or more microprocessors within the IBR.

In accordance with the previous description of the IBR Interface Bridge608, the 802.1 MAC instances of FIGS. 31 and 32 correspond respectivelyto Ethernet interfaces 604 and are substantially conventional. Each802.1 MAC instance is paired with a respective 802.3 PHY instance ofwhich exemplary types include 100 Base-T, 1000 Base-T and variousmembers of the 1000 Base-X family of fiber optic interfaces.

In many exemplary embodiments of the IBR, the IBR Logical Link Control(IBR LLC) layer of FIGS. 31 and 32 is compatible with IEEE 802.2. Inaccordance with the previous description of the IBR Interface Bridge,the IBR LLC layer also uses substantially conventional processes tobridge, for example, 802.2 compatible frames across disparate mediatypes as exemplified by the 802.1, the 802.11 and the IBR MAC layers ofFIGS. 31 and 32. Furthermore, in accordance with the previousdescription of the IBR Interface Bridge, the IBR LLC layer also usessubstantially conventional processes to switch and/or load balance 802.2compatible frames amongst multiple 802.1 MAC instances if available.

FIGS. 31 and 32 depict an exemplary IEEE 802.11 compatible radio (alsoknown as “WiFi”) including 802.11 MAC and 802.11 PHY layers and one ormore antennas. In some IBR embodiments, such an 802.11 radio, which maybe compatible with 802.11b, 802.11a, 802.11g, 802.11n or other 802.11PHY layer variants, within the IBR may be configured as an access point(AP) to allow significant wireless local area network (WLAN) traffic tobe backhauled by the IBR over its wireline or wireless interfaces asappropriate. In some embodiments, the same antennas may be utilized inboth the IBR PHY, which typically includes the IBR Modem 624, IBRChannel MUX 628, IBR RF 632 and IBR Antenna Array 648 in reference toFIGS. 6 and 7, and the 802.11 PHY (e.g., via the IBR RF Switch Fabric ofFIG. 10 (not shown)). Note further that multiple instances of the 802.11radio can also be connected to the IBR LLC in an analogous manner toconnecting multiple Ethernet interfaces.

Another IBR embodiment may provide a more limited 802.11 radiocapability for local configuration purposes (to the exclusion of WLANtraffic and general access). Traditionally PTP and PMP systems haveprovided a “console” input for local configuration via a command lineinterface of radios in the field particularly for situations wherenetwork configuration is either unavailable or undesirable. However,when an IBR is deployed, for example, on a street light, traffic light,building side, or rooftop, a wired console access with a terminal may beextremely inconvenient and/or dangerous. Thus, in some embodiments,802.11 radios are deployed as an access point such that terminalsincluding smartphones, tablets, laptops or other portable computingdevices with 802.11 station capability can connect to such IBRs in sucha console mode. In one embodiment, this deployment is used solely forconfiguration purposes. Configuration by a terminal with 802.11 stationcapability is also possible in the case of one or more 802.11 APsdeployed for WLAN access purposes. Note further that for theconfiguration-only 802.11 AP in an IBR, exemplary embodiments may alsodeploy the 802.11 radio as an Ethernet device on one of the P Ethernetinterfaces depicted in FIGS. 6, 7, 31 and 32, or via some otherIBR-internal wired or bus interface. In such configuration-only 802.11AP IBR embodiments, one or more separate antennas for 802.11 use wouldtypically be provided by the IBR. Furthermore, the configuration-only802.11 interface is not restricted to AP mode only but can also includepeer to peer (“IBSS”), AP mode at the terminal, or WiFi direct.

With reference to FIGS. 31 and 32, typical applications protocols suchas Hyper Text Transfer Protocol (HTTP), Simple Network ManagementProtocol (SNMP), and others can also be implemented using substantiallyconventional software processes or modules executing on one or moremicroprocessors within exemplary IBRs. Such applications protocols canget access to or be accessed from any of the depicted network interfacesin FIGS. 31 and 32 via industry-standard Transmission Control Protocol(TCP) or User Datagram Protocol (UDP) transport layer protocols and theInternet Protocol (IP) network layer protocol. TCP, UDP and IP can beimplemented using substantially conventional software processes ormodules executing on one or more microprocessors within exemplary IBRs.Note that in certain deployments, such as backhaul within a cellularoperator's radio access network (RAN), messages to or from HTTP, SNMP orother applications protocols may need to be sent either via othertransport and network protocols or via encapsulation of TCP/UDP and IPsegments and packets within such other transport and network protocols(not shown in FIGS. 31 and 32). Note further that messages to or fromapplications protocols in the IBR may also bypass the TCP/UDP and IPprotocols, or their equivalents, entirely during a console mode (notshown) using a locally connected terminal.

FIGS. 31 and 32 also illustrate an exemplary element entitled “IBRControl” which is typically implemented as a distribution plane ofinterconnects, buses and/or controllers that arbitrate directcommunications between the RRC and RLC entities with various elements ofthe IBR MAC and IBR PHY. Because the IBR Control does not passinformation amongst such entities by using the communications protocolsstack, such information can be communicated with minimal latency forreal time control.

FIG. 33 illustrates an exemplary implementation of the IBR MAC 612. TheIBR MAC 612 is shown in terms of various functional elements and some oftheir relationships to each other as well as to the IBR PHY, the IBRLLC, the RRC, the RLC and the IBR Control. Operation of the IBR MAC ofFIG. 33 may depend on the type of “superframe” timing utilized.Typically, IBRs send a single PPDU per Modulator Core-j (or per transmitstream k), which may include Training Data blocks as described above, ina given transmit superframe (TxSF) and conversely will receive a singlePPDU per Demodulator Core-j (or per receive stream l) in a given receivesuperframe (RxSF).

As shown in FIG. 33, the IBR MAC 612 includes a IBR MAC managemententity 3304, a MAC Tx Buffer and Scheduler 3308, an IBR MAC ControlEntity 3312, a MAC Rx Buffer 3316, a decryption block 3320, anencryption block 3324, a MPDU header generator 3328, a first FCSgenerator 3332, a MPDU header analyzer 3336, a second FCS generator 3340and a FCS Pass? analyzer 3344.

FIG. 34 illustrates channel activity for a modulator and demodulatorpair j with data interface streams Tx-j and Rx-j respectively versustime for an exemplary IBR using FDD with fixed superframe timing.Similarly, FIG. 35 illustrates analogous channel activity for TDD withfixed superframe timing and FIG. 36 illustrates analogous channelactivity for TDD and Collision Sense Multiple Access (CSMA) withvariable superframe timing.

With reference back to FIG. 33, the operation of the IBR MAC is firstdescribed in the context of an example based on TDD/CSMA with variablesuperframe timing. As shown in FIG. 36, in CSMA transmissions by eitherthe AE-IBR or RE-IBR paired in an active link, whether PTP or PMP,defers to other channel activity detected within its present receiveantenna pattern. In practice, this can be achieved by using the receiveportion of the IBR PHY to determine in channel and in-view signal energywhich if above a threshold causes an inhibit control signal at either aTx PLCP Mod-j of FIG. 27 (not shown) corresponding to the affected linkor at the MAC Tx Buffer and Scheduler of FIG. 33 (not shown) identifyingthe affected link. For purposes of this description, the latter optionis considered. Furthermore, it is assumed that for this specific exampleevery MSDU or frame request at the MAC Tx Buffer and Scheduler resultsin a single MPDU based on such MSDU or frame request (see, for example,FIG. 4) that is then matched to a single PPDU in the Tx PLCPcorresponding to a given link. Other variants are possible with the IBRMAC and IBR PHY as discussed subsequently.

With reference again to FIG. 33, the primary source of MSDUs to the MACTx Buffer and Scheduler 3308 is typically in the data plane from the IBRLLC 3348. Either MSDUs and/or frame requests in the control plane canoriginate from the RRC 660 (or indirectly from the optional IBMS Agent700), the RLC 656, the IBR MAC Management Entity 3304, or the IBR MACControl Entity 3312. Exemplary details of frame requests and/or MSDUsoriginating from the RRC 660 or RLC 656 are discussed subsequently.

Exemplary frames originating at the IBR MAC Management Entity 3304include those associated with management processes such as Association,Authentication and Synchronization. In some cases, the IBR MACManagement Entity 3304 may send an MSDU payload to the MAC Tx Buffer andScheduler 3308 with a specific frame request type. In other cases, theIBR MAC Management Entity 3304 may send only the specific frame requesttype wherein all relevant information to be conveyed to the receivingIBR(s) will be present in the MPDU Header to be generated based on thedetails of the frame request type and other information presently knownto the IBR MAC.

In some embodiments, Association and Authentication processes can occurvia exchange of management frames in a substantially conventionalfashion. A particular RE-IBR may choose to associate with a given AE-IBRby sending an Association Request management frame directed to theAE-IBR based on either advertised information received from such AE-IBRand/or configuration information currently present in the RE-IBR. Uponreceipt of an Association Request, the AE-IBR can proceed according toits presently configured policies to associate with or deny associationto the RE-IBR. If associated, the AE and RE would exchangeauthentication frames in substantially conventional fashion to ensurethat compatible encryption/decryption keys are present for subsequentframe exchanges.

In exemplary IBRs, an RE-IBR can, for example, associate with adifferent AE-IBR if its present AE-IBR (or its wireline link interface)fails, the link throughput falls below a minimum threshold, the presentAE-IBR forces a disassociation, the present AE-IBR inhibits linkresource allocations to the RE-IBR below a minimum threshold, or theRE-IBR becomes aware of a preferred AE-IBR, all as set by certainconfiguration policies at the time as may be set by the optional IBMSAgent 700 as shown in FIG. 33. Furthermore, certain exemplary IBRs witha plurality of modulator cores can as an RE-IBR maintain a plurality ofcurrent associations with multiple AE-IBRs per configuration policies toenable enhanced throughput and/or link diversity.

Another set of exemplary management frames issued by the IBR MACManagement Entity 3304 concerns synchronization, status and presence.Periodically, (as configured or directed by the optional IBMS Agent 700)an exemplary AE-IBR may choose to send a Synchronization Frame thatadvertises certain capabilities of the AE-IBR including wireline linkfailure conditions and provides a time stamp reference usable byexemplary RE-IBRs for timing synchronization as either broadcastuniformly across the full directionality possible for the IBR AntennaArray and/or across all current links. Advantageously, particularly foran AE-IBR with multiple associated RE-IBRs in a PMP configuration, sucha Synchronization Frame (or other such management frame) can direct oneor more RE-IBRs to make internal reference timing offsets such that thetime of arrival of transmissions from such RE-IBRs is more optimallyaligned for best simultaneous reception at the AE-IBR in either FDD orTDD with fixed superframe timing (see FIGS. 34 and 35). For example, theAE-IBR may determine such timing offsets within theAcquisition/Synchronization element of the IBR Channel MUX 628 byanalyzing preamble samples corresponding to particular RE-IBRtransmissions.

With reference again to FIG. 33, exemplary frames originating at the IBRMAC Control Entity 3312 include those associated with control processessuch as Acknowledgement and Access Control. Analogously to the IBR MACManagement Entity 3304, the IBR MAC Control Entity 3312 may send an MSDUpayload to the MAC Tx Buffer and Scheduler 3308 and/or a specific framerequest type wherein relevant information can be conveyed to thereceiving IBR in the MPDU Header.

In exemplary IBRs, Acknowledgement Frames can provide an ACK or NACKindication for configured frame types recently received. The frames areidentified uniquely and then set to ACK or NACK based on the receivepath FCS comparison process illustrated in FIG. 33. To minimizetransport overhead, implementation complexity and transmit latency, someembodiments of the IBR utilize a NACK protocol administered by the IBRMAC Control Entity. With reference to FIGS. 34-36, upon receipt of oneor more MPDUs within RxSF(s), where “s” is a superframe sequence number,then if any MPDU has an FCS failure within RxSF(s), a NACK bit is set inthe MPDU Header of the first (or each) MPDU within TxSF(s+1). If theoriginating IBR detects a positive NACK in TxSF(s+1) at its receiver,then such an IBR will re-transmit via its MAC Tx Buffer and Scheduler3308 the MPDU (or MPDUs) associated with RxSF(s) using link parametersas determined by its RLC 656 at that time. For the FDD case of FIG. 34,such re-transmitted MPDU (or MPDUs) originally from RxSF(s) would atleast partially be received in RxSF(s+2) whereas for the TDD cases ofFIGS. 35 and 36, it would be RxSF(s+1). If there were no FCS failures inRxSF(s), or the originating IBR fails to detect a positive NACK inTxSF(s+1) for any reason, then the originating IBR will clear the MPDU(or MPDUs) associated with RxSF(s) from its MAC Tx Buffer and Scheduler3308. Note that unlike conventional ACK protocols, this exemplary IBRNACK protocol does not guarantee 100% delivery of MPDU. However,occasional MPDU delivery failures are typically correctable at a higherlayer for data plane MSDUs (i.e., using TCP or an equivalent transportlayer protocol), non-catastrophic for management or control plane framesto the IBR MAC Management Entity 3304, the IBR MAC Control Entity 3312,the RRC 660, or the RLC 656, and preferable to the increased latency andjitter of conventional ACK protocols applied to backhaul for those dataplane MSDUs not subject to higher layer correction (i.e. using UDP or anequivalent transport layer protocol). Note further that this NACKprotocol is also advantageously used when PLCP Header FCS failures occurby, for example, having the Rx PLCP Demod-j discarding such a failedPPDU but substituting an MPDU with correct link ID but dummy FCS toRxD-j in FIGS. 29 and 30 to trigger an FCS failure at the IBR MAC 612.

In some embodiments, Access Control Frames are initiated at the IBR MACControl Entity 3312 to control the behavior of other IBRs with currentlinks to the initiating IBR. Such Access Control Frames can, forexample, restrict the rate at which data plane MSDUs are sent, restrictthe timing in which data plane MSDUs are sent, or temporarily inhibitfurther data plane MSDUs from being sent in a local overloadingscenario. For example, an exemplary AE-IBR could utilize Access ControlFrames to enforce access policies discriminatorily amongst multipleRE-IBRs according to certain configuration parameters such as a ServiceLevel Agreement (SLA). Such access policies may also be set via theoptional IBMS Agent 700 as shown in FIG. 33.

With reference to FIG. 33, the MAC Tx Buffer and Scheduler 3308temporarily stores data plane MSDUs from the IBR LLC 3348 as well asframes or frame requests from the IBR MAC Management Entity 3304, IBRMAC Control Entity 3308, RRC 660 and/or RLC 656 until such data can bescheduled for transmission within one or more MPDUs. Such scheduling isadvantageously performed based on policies either configured locally oroptionally communicated from the IBMS Agent 700 via the RRC 660 and theIBR Control 3352. In some embodiments, data plane MSDUs from the IBR LLC3348 usually are the lowest priority in the queue for schedulingcomposition of the next MPDU. If the rate of MSDU delivery from the IBRLLC 3348 exceeds the instant link delivery capability, then to avoidbuffer overflow the MAC Tx Buffer and Scheduler 3308 may have an Xon/offcontrol feedback to the IBR LLC 3348 to cause the IBR LLC 3348 to stopsending MSDUs until the appropriate buffer status returns. This mayresult in the IBR LLC 3348 causing certain other interfaces such as anEthernet or WiFi interface to reduce MSDU traffic. To the extent thatthe ratio of control plane traffic to data plane traffic is small, as isby design for some embodiments, then scheduling priority amongst framesother than MSDUs from the IBR LLC 3348 is unimportant and “first-in,first-out” is sufficient.

Note that the foregoing embodiment of the MAC Tx Buffer and Scheduler3308 performs scheduling based on frame type without regard for otherinformation such as the PCP within an 802.1 MAC frame as may be presentin MSDUs from the IBR LLC 3348. The IBR LLC 3348 in its bridgingcapacity may forward MSDUs to the MAC Tx Buffer and Scheduler 3308 inorder based on PCP (or other such QoS or CoS indicators) so that the MACTx Buffer and Scheduler 3308 need not repeat the queuing exercise. Inalternative embodiments, such QoS prioritization of MSDUs can also beperformed at the MAC Tx Buffer and Scheduler 3308 instead.

Upon scheduling an MPDU for transmission as described above, the MAC TxBuffer and Scheduler 3308 causes the MPDU Header Generator 3328 tocompose an MPDU Header specific to the pending MPDU. In conventionalIEEE 802 based communications systems, such an MPDU Header would includeat least the physical address (also known as the “MAC address” or the“hardware address”) of both the origination and destination IBRs.However, sending such physical addresses (typically 48 or, morerecently, 64 bits) in every MPDU unduly burdens the IBR links withunnecessary overhead that reduces MAC efficiency. Thus, in someembodiments, a Link Identifier (LID) is substituted in every MPDU headerinstead. Exemplary LID implementations can be as few as 16 bits inlength. For example, each exemplary LID may include an AE-IBR identifierof 10 bits and an identifier of 6 bits for an RE-IBR presentlyassociated with the AE-IBR. This is possible because in some embodimentsthe IBRs are configured in view of their fixed geographic positions inthe field as set at time of deployment or optionally controlled by theIBMS Agent 700 such that no overlapping AE-IBR identifiers are withinradio range of RE-IBRs possibly associated with them. The AE-IBR mayassign a locally unique RE-IBR association identifier field as part ofthe association process. For unassigned links, all zeros (or ones) canbe sent for LID and then the frame payload, for example a managementframe used in the association process, can include the full physicaladdresses as appropriate. Note that even if, in the alternative, thatlonger (possibly 24 bits or 32 bits) “regionally-unique” or even“globally-unique” LIDs were used, then because the overall number ofworldwide backhaul links is generally much less the overall number ofworldwide network devices, such extended length LIDs can still be muchshorter than traditional IEEE 802 based addressing schemes.

A Frame Type Identifier (FTI) may be placed in the MPDU header by theMPDU Header Generator 3328. In one embodiment, the FTI is no more than 5bits total. In one particular embodiment, the FTI is part of a FrameControl Field (FCF) of 8 bits, and the other 3 bits include 1 bit forthe NACK protocol control bit indicator (set to 1 if the previousRxSF(s) had an MPDU with an FCS failure), 1 bit to indicate if theinstant frame of the MPDU payload is encrypted, and 1 bit to indicate ifthe instant frame of the MPDU payload is the last frame (LF) in thepayload. Alternatively, the 1 bit for the NACK can be 1 bit for the ACKindicator (set to 1 if the previous RxSF(s) had an MPDU without an FCSfailure) if an ACK protocol is used for the instant MPDU FTI. Followingthis FCF byte, a 16 bit Fragment and Length Indicator (FLI) is placedsequentially in the MPDU header, wherein, for example, 3 bits of the FLIindicates by 1 bit if the instant frame payload is the last fragment andby 2 bits the fragment sequence number and 13 bits indicate the instantframe payload length in bytes. Following the 2 FLI bytes, an 8 bit FrameSequence Number (FSN) is placed sequentially in the MPDU header. The FSNare typically sequentially generated except where repeated for thoseframe payloads sent as fragments. If LF=1 in the initial FCF byte of theMPDU header (as would be the case for the single MSDU of frame payloadper MPDU scenario described above for TDD/CSMA), then the MPDU header iscomplete. If the MAC Tx Buffer and Scheduler 3308 is configured topermit concatenation of MSDUs or other frame payloads up to some maximumMPDU payload (as would be compatible with many TDD/CSMA deployments),then LF=0 when FLI describes an instant frame payload lengthsufficiently low to allow another available frame payload to beconcatenated within the maximum MPDU payload length and an additionalFCF, FLI and FSN combination would be generated at the MPDU HeaderGenerator 3328 and repeated until an FCF with LF=1 is encountered. Notethis process of concatenated FCF, FLI and FSN fields within an MPDUheader corresponding to concatenated frame payload can also beadvantageously applied to fixed superframe timing in either FDD or TDDas illustrated in FIGS. 34 and 35.

The MAC Tx Buffer and Scheduler 3308 further provides the one or moreframe payloads that form the MPDU payload to the Encryption element 3324to be encrypted on a frame payload by frame payload basis as indicatedin the FCF using substantially conventional encryption algorithms. Anexemplary algorithm suitable for the IBR is the Advanced EncryptionStandard (AES) which has been published by the National Institute ofStandards and Technology. In one embodiment, the IBRs use AES with a 256bit key length. Other key lengths and other encryption algorithms arealso possible for exemplary IBRs. Exemplary IBRs can also employencryption for all frames after encryption keys are exchanged duringauthentication (and even including association and authentication framesto the extent encryption keys sufficient at least for association andauthentication are provided to IBRs via, for example, factory setting orconsole mode interface).

The encrypted (and/or unencrypted as desired) frame payload(s) and theMPDU header are then concatenated together as shown in FIG. 33 and thenpassed through an FCS Generator 3332 that generates an FCS (e.g., of atleast 32 bits in length). Alternatively, some IBR embodiments may alsoencrypt (and decrypt) MPDU headers. Other IBR embodiments may attainsome measure of MPDU header privacy (and PLCP header privacy) by settingscrambler and descrambler parameters (see FIGS. 19-22) derived from, forexample, the encryption keys and/or LID. The FCS is then appendedfollowing the MPDU header and frame payload(s) as also shown in FIG. 33to complete the composition of an MPDU to be sent to the IBR PHY 3356over the Tx Data interface.

Note also that for those frame types, particularly certain managementframes originating at the AE-IBR of a PMP configured deployment, thatare intended to be broadcast to all current links at an IBR, such framepayloads may be distributed to all such links in parallel at the MAC TxBuffer and Scheduler 3312 such that the same frame payload is providedto at least one MPDU corresponding to each current link. The IBRstypically generate very little broadcast traffic and most Ethernet orWiFi broadcast traffic on the other IBR interfaces is filtered at theIBR LLC 3348.

With reference to FIG. 33, the receive path of the exemplary IBR MACperforms essentially the reverse operations described above in thetransmit path. Starting with the IBR PHY 3356, after the Rx PLCPanalyzer 1808 (see FIG. 18) recovers an MPDU, it is passed via the RxData interface in FIG. 33 to a splitter 3360 within the IBR MAC. Thesplitter 3360 removes the trailing bits of the MPDU corresponding to theFCS, or RxFCS. The remainder of the MPDU minus RxFCS is passed throughan FCS Generator 3340, the result of which is compared at FCS Pass?analyzer 3344 to determine if it is identical to RxFCS. If it isidentical, then a known-good (barring the extraordinarily rare falsepositive FCS event) MPDU is received. The FCS Pass? status iscommunicated to the IBR MAC Control Entity 3312, the MAC Rx Buffer 3316,the MPDU Header Analyzer 3336, and the RRC 660 and RLC 656 (foranalytical purposes) as shown in FIG. 33. The MPDU Header Analyzer 3336receives from a second splitter the leading bits of the MPDUcorresponding to a minimum length MPDU header described above. If LF=1,then the remaining payload portion of the MPDU is passed to theDecryption element 3320 (or bypassed if appropriate) for decryption whenthe encryption indicator bit of the FCF is set. The MPDU Header Analyzer3336 may also verify (or cause to be verified) that the payload lengthcorresponds to that described in the FLI field of the MPDU Header andthat the LID is valid. Upon decryption (as appropriate), the MPDUpayload is passed to the MAC Rx Buffer 3316 which then forwards the MSDUor frame payload to the IBR MAC Management Entity 3304, the IBR MACControl Entity 3312, the IBR LLC 3348, the RRC 660 or the RLC 656 asappropriately directed by the MPDU Header Analyzer 3336 based on LID andFTI. The MPDU Header Analyzer 3336 may also directly signal the IBR MACControl Entity 3312 with the status of the NACK bit in the exemplaryMPDU header described above. In the event that the instant frame payloadis a fragment, as evident by the FLI field in the exemplary MPDU headerdescribed above, then the MPDU Header Analyzer 3336 instructs the MAC RxBuffer 3316 based on the FSN field to either store the fragment,concatenate the fragment to one or more other stored fragments, ordeliver the multiple fragments if the instant fragment has the lastfragment indicator within the FLI field as described above. Optionallyif a positive ACK protocol is used for 100% delivery of every MPDU(versus the superframe NACK protocol described above), then the MDPUHeader Analyzer 3336 also verifies the FSN for duplicated detection anddiscards (or causes to be discarded) such duplicate payloads.

In the event that the MPDU being received has multiple concatenatedMSDUs and/or frame payloads as described optionally above, then the MPDUHeader Analyzer 3336 interacts with the second splitter via the feedbacksignal shown to continue parsing off consecutive MDPU header bytes, forexample the repeated exemplary FCF, FLI and FSN fields, until an FCFwith LF=1 is encountered. The above described process for the MPDUHeader Analyzer 3336 directing the Decryption element 3320 and the MACRx Buffer 3316 to deliver the multiple payloads is then repeated eitherserially or in parallel as desired until all contents of the MPDUpayload have been resolved by the IBR MAC.

In the event that the FCS Pass? analyzer 3344 described above determinesan FCS failure then the receive path of the exemplary IBR MAC operatesdifferently depending on certain options that may be designed into theIBR or selected based on configuration policies. For the superframe NACKprotocol described above, the FCS failure is directly communicated tothe RRC 660 and RLC 656 (via the IBR Control 3352 of FIGS. 31-33) fortheir analytical purposes and is directly communicated to the IBR MACControl Entity 3312 so that TxSF(s+1) is sent with NACK=1 in theexemplary MPDU header described above. Optionally, the MPDU HeaderAnalyzer 3336 may attempt to determine if an apparently valid MPDUheader is present, and, if so, attempt to deliver certain MSDUs or framepayloads as described above for the case of a known-good MPDU. However,as directed by configuration policies, certain frame types may bediscarded. For example, in one embodiment, only MSDUs to the IBR LLC3348 that correspond to apparently valid MPDU header fields would bedelivered in an FCS failure situation, and even then only afterbuffering such MSDUs for one additional superframe period to determineif a duplicate FSN MSDU may be available due to the re-transmissionattempt associated with the superframe NACK protocol described above.Alternatively, in the case of an ACK protocol that guarantees 100%delivery of known-good MPDUs, then the entire MPDU payload is discarded.

For the types of data throughput rates and superframe lengths that arepractical, the single payload MPDU per superframe example describedabove for the TDD/CSMA example of FIG. 36 is not an optimum choice forthe fixed superframe timing examples for FDD in FIG. 34 or TDD in FIG.35. One alternative uses concatenated multi-payload frame MPDUs up tosome maximum length and then forwards such MPDUs to the Tx PLCP tocompose PPDUs as described previously. Another alternative, compatiblewith the superframe NACK protocol described above, would have the MAC TxBuffer and Scheduler 3308 utilize the same information from the RLC 656and the RRC 660 available to the Tx PLCP and thus compose a singlemulti-payload MPDU that will, with appropriate padding added in the TxPLCP to an integer block count, maximally occupy the available PPDU forthe pending superframe.

Note that for the TDD case depicted in FIG. 35 with fixed superframetiming, IBRs are capable of operating in such modes wherein thesuperframe timing in the forward link from AE to RE may be differentthan in the reverse link. For a PMP AE-IBR this typically requires thatall RE-IBRs adopt the same superframe timing parameters in associationwith this particular AE-IBR. Some RE-IBRs, in either PTP or PMPconfigurations, may be simultaneously associated with two or moreAE-IBRs to the extent such IBRs permit this policy and have sufficientradio resources to maintain such links. Given that multiple AE-IBRsassociated with such RE-IBRs can adopt different TDD fixed superframetiming parameters relative to each other, such timing parameters may becommunicated to all RE-IBRs via management frames originating in the IBRMAC Management Entity 3304. Similarly, to the extent an RE-IBR islimited in superframe timing access due to such multiple associations,such an RE-IBR can communicate such restrictions to the AE-IBRs viamanagement frames from its IBR MAC Management Entity 3304.

For PMP configurations, the AE-IBR can advantageously transmit to orreceive from multiple RE-IBRs simultaneously in a given radio channelusing SDMA as described above. To the extent that the number of RE-IBRsassociated with an AE-IBR exceeds either the AE-IBR's available SDMAresources or certain RE-IBRs are not spatially separable by the AE-IBR,then simultaneous transmissions to such RE-IBRs would require multipleradio channels which in practice is often either undesirable orimpractical. Another alternative for maintaining links to RE-IBRswherein such SDMA capabilities are being exceeded is through the use ofTime-Division Multiplexing (TDM). For either an FDD or TDD system withfixed superframe timing, such as depicted in FIGS. 34 and 35, an AE-IBRcan use TDM to designate certain “subframes” within a given superframeapplied to a modulator/demodulator resource pair j to particular LIDs.Management frames from the AE-IBR's IBR MAC Management Entity 3304 can,in exemplary embodiments, inform affected RE-IBRs of such LID to TDMsub-frame mappings and associated subframe timing parameters. In someembodiments, any RE-IBR associated with two or more AE-IBRssimultaneously could request that such LID to TDM sub-frame mappingsand/or timing parameters account for its local timing preferences foroptimally maintaining such simultaneous links. The MAC Tx Buffer andScheduler 3308 may compose MPDUs in view of TDM sub-frame timingparameters that are assigned such that the TxPLCP can optimally utilizean integer number of transmit blocks in each sub-frame. The number ofsub-frames per superframe are typically relatively low (e.g., usuallyless than 5).

To the extent that IBRs deployed in the field are within co-channelinterference range of each other and configured to use overlappingchannels in general or periodically, such IBRs advantageouslysynchronize their TDD fixed superframe timing to minimize simultaneousco-channel TxSF/RxSF overlaps amongst disparate links. Several exemplarysynchronization strategies may be used by such IBRs to align superframetiming boundaries in such scenarios. For example, if “free-running” or“self-synchronizing” IBRs are able to detect at least a preamble, atraining block, a PLCP header, an unencrypted MPDU header, or otherinformation such as management frames with timing information that maybe decipherable without link-specific encryption keys that correspond tolinks involving a different AE-IBR, then the slower in time IBRs mayadopt the superframe timing boundary and cadence of the faster in timeIBRs. At the AE-IBR, which may act as a local timing master, this can beperformed directly by making a timing offset and communicating it to isassociated RE-IBR(s). At the RE-IBR, which may be able to detectdisparate link information otherwise undetectable at its associatedAE-IBR, the RE-IBR can inform its AE-IBR via a management frame of anytiming offset necessary to obtain local disparate AE-IBR co-channelsuperframe timing synchronization. It will be appreciated that thisprocess is ongoing because after synchronizing, the reference clocks inthe AE-IBRs inevitably will drift differently over time.

In certain field deployment scenarios, IBRs located in the same regionalarea may be capable of undesirably interfering with each other at rangesbeyond their ability to detect and synchronize as described above. Analternative synchronization strategy better suited to this situationwould utilize a network-wide central synchronization capability. Oneexample of this would be the use of Global Positioning Satellite (GPS)timing at each AE-IBR. GPS is more than adequate in terms of timingaccuracy for the needs of synchronizing superframe timing boundaries.However, GPS adds cost, size, power consumption and form factorconstraints that may be undesirable or unacceptable to some IBRs. Notefurther that because IBRs are designed to operate in street leveldeployments with obstructed LOS, GPS may fail to operate in places whereRE-IBRs function normally. Another alternative would be to use asystem-wide synchronization technique such as SynchE or IEEE 1588v2. Inthis scenario, AE-IBRs are configured to derive timing parameters in aconsistent fashion. Alternatively, the AE-IBRS include IBMS Agentscapable of coordinating such configurations when co-channel operation ina mutual interfering deployment is encountered.

In the deployment scenario where multiple AE-IBRs are co-located (e.g.,at a single pole, tower, building side or rooftop), even if such IBRsare configured to avoid co-channel operation, at least some form oflocal superframe timing synchronization for TDD may be utilized to avoidoverloading receiver RF/analog circuits from simultaneous in-bandtransmissions and receptions amongst the co-located IBRs. One exemplarystrategy for the above synchronization would be to distribute by hardwiring a local superframe timing synchronization signal which can beconfigured at or arbitrarily assigned to any one of the co-located IBRs.

Note that the foregoing descriptions and figures for exemplary IBRembodiments have provided minimal internal details on the distributionof various clocks and timing references amongst the structural andfunctional elements of the IBR. These exemplary embodiments can all berealized using substantially conventional strategies for internal timingdistribution.

As described above, IBRs with fixed superframe timing may use a NACKprotocol wherein a previous superframe FCS failure in receive causesNACK=1 in an MPDU header of the respective link in transmitting its nextsequential superframe PPDU back to the sender of the MPDU received inerror. If the original sender detects NACK=1, then a re-transmission ofthe previous superframe PPDU contents occurs at the direction of its MACTx Buffer and Scheduler 3308. Otherwise, the original sender discardsthe previous superframe PPDU contents at its MAC Tx Buffer and Scheduler3308. This approach is different from many conventional wireless datanetworking protocols that use ACK receipt to guarantee 100% delivery atthe MAC layer even if theoretically unbounded re-try attempts may berequired. This fixed superframe NACK protocol is similar to using an ACKprotocol with a “time to live” of effectively one superframe durationafter initial transmission. This approach advantageously bounds thelatency at a very low length for processing frames through the IBR MACwithout resorting to reliability of simply the raw frame error rate. Byallowing one immediate re-transmission opportunity, this fixedsuperframe NACK protocol effectively produces a net frame error ratethat is the square of the raw frame error rate. For example, if the rawframe error rate were 10⁻³ (1 in 1000), the net frame error rate per thefixed superframe NACK protocol should be approximately 10⁻⁶ (1 in1,000,000).

To improve the reliability of the fixed superframe timing NACK protocoleven further, IBRs may have the TxPLCP set a 1 bit field in the PLCPheader(s) for the Modulator Core j corresponding to the LID with NACK=1(or with a previous RxSF(s) PLCP header FCS failure as described above).This approach advantageously exploits the fact that PLCP headers,typically sent at the most reliable MCS, are of short duration andalways sent immediately after the Training Block 0 (see FIG. 26) wherethe channel equalization coefficients most accurately reflect currentconditions. Thus, a NACK bit sent in a PLCP header is likely to bereceived accurately even if the PPDU payload that includes the MPDUheader NACK field is not. In the case where NACK=1 in a PLCP header, anexemplary PLCP Header Analyzer 3004 (see FIG. 30) will signal theexemplary MAC Tx Buffer and Scheduler 3308 via the IBR Control 3352. Inthe case where an originator of an MPDU does not receive a valid NACKfield in either the PLCP header or the next MPDU header of a subsequenttransmit superframe, then the MAC Tx Buffer and Scheduler 3308 maychoose to re-send the pending MPDU(s), possibly at a more reliable MCSas directed by the RLC 656, or may choose to discard the pending MPDU(s)so as not to cascade latency for further frames awaiting transmission.Such a decision may be made based on configurable policies that may varywith current conditions or be updated by the IBMS Agent 700. To theextent that a “blind” retransmission is made due to an invalid NACKfield wherein the actual NACK field value was NACK=0, then the receivepath MPDU Header Analyzer 3336 at the destination IBR will discard there-sent MPDU(s) based on duplicate detection of the FSN at the MPDUHeader Analyzer 3336 shown in FIG. 33. The MPDU Header Analyzer 3336 mayalso report incidents of duplicate detections of MPDUs for a given LIDto the RRC 660 and RLC 656 as shown in FIG. 33. This provides anadvantageous additional local indication to the RRC 660 and RLC 656 ofproblems being encountered in the transmit direction from such an IBRfor the particular LID.

With reference to FIGS. 31 and 32, the RRC 660 and RLC 656 interact withthe IBR MAC 612 and various elements of the IBR PHY both via “normal”frame transfers and direct control signals via the conceptual IBRControl plane. Both the RRC 660 and the RLC 656 may execute concurrentcontrol loops with the respective goals of optimizing radio resourceallocations and optimizing radio link parameters for current resourcesin view of the dynamic propagation environment conditions, IBR loading,and possibly system-wide performance goals (via the optional IBMS Agent700). It is instructive to view the RLC 656 as an “inner loop”optimizing performance to current policies and radio resourceallocations for each active link and to view the RRC 660 as an “outerloop” determining if different policies or radio resource allocationsare desirable to meet overall performance goals for all IBRs currentlyinteracting with each other (intentionally or otherwise). Typically boththe RRC 660 and the RLC 656 are implemented as software modulesexecuting on one or more processors.

The primary responsibility of the RLC 656 in exemplary IBRs is to set orcause to be set the current transmit MCS and output power for eachactive link. In one exemplary embodiment described above, the RLC 656provides information to the TxPLCP that enables, for example, a PLCPController 2812 in FIG. 28 to set the MCS for a particular LID. Inanother embodiment compatible with the single MPDU for fixed superframetiming with NACK protocol example of IBR MAC operation, the RLC 656determines the MCS for each LID and then communicates it directly to theMAC Tx Buffer and Scheduler 3308 of FIG. 33 and the PLCP Controller 2812for every TxPLCP Mod-j of FIGS. 27 and 28. The RLC 656 causes thetransmit power of the IBR to be controlled both in a relative senseamongst active links, particularly of interest for the AE-IBR in a PMPconfiguration, and also in an overall sense across all transmits chainsand antennas. In an exemplary IBR embodiment, the RLC 656 can cause suchtransmit power control (TPC) as described above by setting parameters atthe Channel Equalizer Coefficients Generator 2332 of FIG. 23 (forrelative power of different simultaneous modulation streams), at eachactive transmit RF chain Tx-m 636 of FIGS. 6, 7 and 16 (for relativepower of different simultaneous RF chains) and at each Front-end PA1104, 1204 within the IBR Antenna Array of FIGS. 6, 7, 10-12 (for totalpower from all antennas).

In some embodiments, the RLC 656 can determine its MCS and TPCselections across active links based on information from various sourceswithin the IBR. For example, the IBR MAC can deliver RLC control framesfrom other IBRs with information from such other IBRs (for example,RSSI, decoder metrics, FCS failure rates, etc.) that is useful insetting MCS and TPC at the transmitting IBR. Additionally, such RLCcontrol frames from an associated IBR may directly request or demandthat the RLC in the instant IBR change its MCS and/or TPC values fortransmit directly on either a relative or absolute basis. For TDD IBRdeployments, symmetry of the propagation environment (in the absence ofinterference from devices other than associated IBRs) makes receiverinformation useful not only for sending RLC control frames to thetransmitting IBR but also for use within the receiving IBR to set itstransmitter MCS and/or TPC. For example, the FCS Pass? analyzer 3344 andMPDU Header Analyzer 3336 of the exemplary IBR MAC in FIG. 33 can supplythe RLC 656 with useful information such as FCS failures, duplicatedetections and ACK or NACK field values. Similarly, each PLCP HeaderAnalyzer 3004 as shown in FIG. 30 can send the RLC FCS failure status onthe PLCP Header and/or PPDU payload(s). Another possibility is to havethe PLCP Header carry a closed-loop TPC control field requesting a TPCstep up or down or stay the same. Additionally, the Channel EqualizerCoefficients Generator 2332 in computing channel weights can provide SNRand/or SINR information per demodulator stream that can help the RLC setMCS and/or TPC as shown in FIG. 23. Each decoder within each demodulatorcore j can provide the RLC 656 with decoder metrics useful for MCSand/or TPC as indicated by FIGS. 20 and 22 for example. And each receivechain Rx-n 640 of FIG. 17 can provide receive signal strength indicator(RSSI) values helpful especially for determining TPC requests back tothe transmitting IBR.

The actual MCS values are typically selected from a finite length listof modulation types and coding rates. Exemplary IBRs can use QAM rangingfrom 2-QAM (better known as BPSK), through 4-QAM (better known as QPSK),16-QAM, 64-QAM, 256-QAM and 1024-QAM. Exemplary IBRs can use a basecoding rate of ⅓ or ½ and then can use “puncturing” (whereinpredetermined bit positions are deleted in transmit and replaced bydummy bits in receive) to derive a set of effective coding rates of, forexample only, ½, ⅔, ¾, ⅚, ⅞, and 9/10. In typical embodiments, thelowest MCS index corresponds to the lowest available QAM constellationsize and the lowest available coding rate (i.e. the most reliabletransmission mode) and the highest MCS index corresponds to theconverse.

The TPC absolute range tends to be lower for IBRs than that desired formany conventional wireless networking systems operating in obstructedLOS due to the more limited range of separations between AE-IBRs andRE-IBRs for backhaul applications (i.e. backhaul radios are almost neverplaced in close proximity to each other). The relative variation indesired power between active links at an AE-IBR may also limited inrange particularly by the transmit DACs.

Many possible algorithms are known for generally relating informationprovided to the RLC 656 as described above to selecting MCS and TPCvalues. In dynamic propagation environments, averaging or dampeningeffects between channel quality information and MCS changes areadvantageously utilized to avoid unnecessarily frequent shifts in MCS.To the extent that an IBR is operating at below the maximum allowableTPC value, it is generally advantageous to permit TPC to vary morequickly from superframe to superframe than the MCS. However, ifoperating at maximum allowable TPC, then it is often advisable toimmediately down select MCS to a more reliable setting upon detection ofan MPDU FCS failure and/or a NACK=1 condition. Conversely, up selectingMCS is usually performed only after repeated superframe metricsindicating a high likelihood of supporting an MCS index increase. At thelimit, where RLC 656 has reached maximum TPC and minimum MCS (mostreliable mode), to maintain ongoing link reliability, the imperativeincreases for the RRC 660 to allocate different resources to enable theRLC 656 to operate again in MCS and TPC ranges with margin for temporalchannel impairment.

The primary responsibility of the RRC 660 is to set or cause to be setat least the one or more active RF carrier frequencies, the one or moreactive channel bandwidths, the choice of transmit and receive channelequalization and multiplexing strategies, the configuration andassignment of one or more modulated streams amongst one of moremodulator cores, the number of active transmit and receive RF chains,and the selection of certain antenna elements and their mappings to thevarious RF chains. Optionally, the RRC may also set or cause to be setthe superframe timing, the cyclic prefix length, and/or the criteria bywhich blocks of Training Pilots are inserted. The RRC 660 allocatesportions of the IBR operational resources, including time multiplexingof currently selected resources, to the task of testing certain linksbetween an AE-IBR and one or more RE-IBRs. The RRC 660 evaluates suchtests by monitoring at least the same link quality metrics as used bythe RLC 656 as evident in the exemplary embodiments depicted in FIGS. 6,7, 17, 18, 20, 22, 23, 29, 30, 31, 32 and 33. Additionally, in someembodiments, additional RRC-specific link testing metrics such as thoseproduced by the exemplary Training Data Analyzer described above forFIG. 30 in relation to the exemplary Training Data Library of FIG. 28are used. The RRC 660 can also exchange control frames with a peer RRCat the other end of an instant link to, for example, provide certainlink testing metrics or request or direct the peer RRC to obtain linkspecific testing metrics at the other end of the instant link forcommunication back to RRC 660.

In some embodiments, the RRC 660 causes changes to current resourceassignments in response to tested alternatives based on policies thatare configured in the IBR and/or set by the optional IBMS Agent 700 asdepicted for example in FIGS. 7, 32, and 33. An exemplary policyincludes selecting resources based on link quality metrics predicted toallow the highest throughput MCS settings at lowest TPC value.Additional exemplary policies may factor in minimizing interference bythe instant link to other AE-IBR to RE-IBR links (or other radio channelusers such as conventional PTP radios) either detected at the instantIBRs or known to exist at certain physical locations nearby as set inconfiguration tables or communicated by the optional IBMS Agent 700.Such policies may also be weighted proportionately to reach a blendedoptimum choice amongst policy goals or ranked sequentially inimportance. Also as discussed above regarding the RLC 656, the RRC 660may have policies regarding the fraction of IBR resources to be used forRRC test purposes (as opposed to actual IBR backhaul operations) thatfactor the current RLC settings or trajectory of settings.

The RRC 660 in one IBR can communicate with the RRC in its counterpartIBR for a particular link AE-IBR to RE-IBR combination. For example, theRRC 660 sends RRC control frames as discussed for the exemplary IBR MACof FIG. 33 or invokes a training mode within the exemplary Tx PLCP asdiscussed for FIG. 28.

In some embodiments, for either PTP or PMP deployment configurations,the selection of either the one or more active RF carrier frequenciesused by the RF chains of the IBR RF, the one or more active channelbandwidths used by the IBR MAC, IBR Modem, IBR Channel MUX and IBR RF,the superframe timing, the cyclic prefix length, or the insertion policyfor blocks of Training Pilots is determined at the AE-IBR for any givenlink. The RE-IBR in such an arrangement can request, for example, an RFcarrier frequency or channel bandwidth change by the AE-IBR by sendingan RRC control frame in response to current link conditions at theRE-IBR and its current RRC policies. Whether in response to such arequest from the RE-IBR or due to its own view of current linkconditions and its own RRC policies, an AE-IBR sends the affectedRE-IBRs an RRC control frame specifying at least the parameters for thenew RF frequency and/or channel bandwidth of the affected links as wellas a proposed time, such as a certain superframe sequence index, atwhich the change-over will occur (or alternatively, denies the request).The AE-IBR then makes the specified change after receiving confirmationRRC control frames from the affected RE-IBRs or sends a cancellation RRCcontrol frame if such confirmations are not received before thescheduled change. In some deployment situations, the RRC policycondition causing the change in the RF carrier frequency and/or channelbandwidth for a particular LID may be a directive from the IBMS Agent700.

The selection of other enumerated resources listed above at an IBR cangenerally be made at any time by any given IBR of a link. Additionally,requests from the opposite IBR can also be made at any time via RRCcontrol frames. An RE-IBR typically attempts to utilize all availablemodulator and demodulator cores and streams as well as all available RFchains to maximize the robustness of its link to a particular AE-IBR. Inan RE-IBR embodiment where at least some redundancy in antenna elementsamongst space, directionality, orientation, polarization and/or RF chainmapping is desirable, the primary local RRC decision is then to chooseamongst these various antenna options. An exemplary strategy forselecting antenna options (or other enumerated resources listedpreviously) at the RE-IBR is to apply alternative selections of suchresources to the Training Data portion of PPDUs described above inrelation to the Tx PLCP and FIGS. 27 and 28. The RRC 660 at the RE-IBRthen compares link quality metrics such as those used by the RLC 656 andtraining data metrics from the exemplary Training Data Analyzer of FIG.30 to determine if the alternatively-selected resources are likely toresult in improved performance based on current IBR RRC policiescompared to the known performance of the currently-selected resources ofthe instant link. If the RRC 660, based on one or more such alternativeselection tests, decides that the instant link's performance goals arelikely to improve by using the alternately-selected resources, then theRRC 660 at such RE-IBR can cause such selections to become the newcurrent settings at any time and without requiring notification to theAE-IBR.

For the RE-IBR alternate resource selection process described aboveapplied to a TDD configuration, channel propagation symmetry for a givenlink (if interference is ignored) may make changing to a correspondingset of resources for transmit from such RE-IBR as have beenalternatively-selected for receive preferable. However, this isgenerally not true for an FDD configuration or a scenario where unknowninterference represents a significant channel impairment or where a PMPAE-IBR has simultaneous links to other RE-IBRs. In such scenarios, anRE-IBR may notify the AE-IBR when such RE-IBR is testingalternately-selected resources in a portion of its Tx PPDUs, whether inresponse to an RRC control frame request by the AE-IBR or by suchRE-IBR's own initiative, and then receive a return RRC control framefrom the AE-IBR that either reports the measured link quality metricsobserved at the AE-IBR and/or directs the RE-IBR to adopt thealternatively-selected resources for future Tx PPDUs from the RE-IBR onsuch link.

For the PTP configuration, an AE-IBR performs its RRC-directed alternateresource selections using substantially the same processes describedabove for the RE-IBR but with the roles reversed appropriately. In thePMP configuration, an AE-IBR may utilize similar RRC-directed testing ofalternate resource selections across its multiple current links but tothe extent that such links depend concurrently on certain resources, thedecision to actually change to different resources may be based onpolicies applied to the benefit of all current links. One strategy forPMP operation of IBRs is to use the maximum possible RF chain andantenna element resources at all times at the AE-IBR and then optimizeselectable resources at the RE-IBRs to best achieve RRC policy goals.

Note that in some deployment situations, spectrum regulations, such asthose set by the Federal Communications Commission (FCC) in the USA, mayrequire active detection of and avoidance of interference with otherusers of the spectrum (such as, for example, radar systems). The processof detecting such other co-channel spectrum users and then changing RFcarrier frequencies to another channel void of such other uses iscommonly called Dynamic Frequency Selection (DFS). Spectrum regulationsmay require that a DFS capability operate in a certain manner inresponse to certain interference “signatures” that may be detected atthe receiver of a certified radio for such spectrum. For example, someregulations require that upon detection of certain pulse lengths andreceived powers that certified radios change immediately to anotherchannel known from some minimum observation time not to have suchinterferers in operation. In some exemplary IBR implementations, suchobservations of alternative channels in advance of needing to make achange can be performed by time division multiplexing certain RF chainand antenna resources to make such measurements using RSSI and/orchannel equalization metrics reported to the RRC 660. In someembodiments, the AE-IBR and the one of more associated RE-IBRscoordinate such observations using RRC control frames to minimallydisrupt backhaul operations and maximally increase the aggregateobservation time and improve the observation accuracy. In otherexemplary IBR embodiments, at least one IBR, typically the AE-IBR, hasat least one Rx-n chain, one antenna and possibly one demodulator corepath through the IBR Channel MUX dedicated to such spectralobservations.

In embodiments with the optional IBMS Agent 700, the above channelobservation techniques of the RRC 660 can also be used in a “probe inspace” mode of operation, either at one IBR or coordinated amongstmultiple IBRs, to observe and record RF channel activity withindesignated portions of the addressable bands of operation. Such spectralanalysis information may be passed from the RRC 660 to the IBMS Agent700 for further analysis and possibly communication to other IBMS Agentsor to other databases or expert systems elsewhere within the IBRoperator's private network or to an external network such as theInternet.

Note also that the DFS operation described above is desirable forexemplary IBRs operating in spectrum bands that do not require DFSexplicitly. For example, if IBRs are deployed in licensed spectrum whereconventional PTP links operate, such conventional links generally lackthe RF carrier frequency agility and radio resource control intelligenceof the IBR. Even if the interference immunity capabilities of the IBRthrough the advantageous combinations of elements described herein issufficient to reject the interference caused by such conventional PTPlinks at the IBR receiver, it is still desirable to have the RRC 660perform DFS to avoid the converse scenario where the IBRs areinterfering with the conventional PTP links. This may be advantageousbecause it minimizes licensed band user conflicts especially if adifferent operator uses the conventional PTP equipment from thatoperating the IBRs. The presence of the conventional PTP link may bedetected in normal operation of the one or more IBRs using a particularRF carrier frequency channel or may be communicated to the RRC 660 viathe optional IBMS Agent 700 that has gathered the information fromanother source. An exemplary technique that the RRC 660 can use in anIBR where N>L and the instant SINR is approximately the same as the SNR(i.e. no significant co-channel interference) per metrics available tothe RRC 660 from the Channel Equalizer Coefficients Generator 2332 is toassign up to N minus L combinations of an antenna element 652, an Rx-nchain, and a channel MUX receive path to a Complex DFT-n to differentfrequency channels than the instant link channel to perform DFS or“probe in space” measurements and spectral analysis. For FDDconfigurations, assigning these combinations to monitor the instant orcandidate transmit frequency channels (possibly during a time when thetransmitter is otherwise inhibited) can allow the RRC to evaluatepotential interference to other conventional PTP links and to adjusttransmit resources accordingly. To the extent that the remaining atleast L receive chains provide sufficient SNR or SINR to maintain theinstant traffic load, this approach allows the RRC 660 to utilizeavailable IBR resources simultaneously for both supporting link trafficand supporting DFS or “probe in space” measurements and spectralanalysis.

As described previously, exemplary IBRs advantageously exploit thepropagation path diversity usually present in an obstructed LOSenvironment to send multiple modulated streams concurrently and thusincrease overall link throughput. For practical reasons regarding actualfield deployments, it is likely that some IBRs will be deployed inlocations where propagation may be dominated at least at some times byunobstructed LOS conditions. In such situations, IBR embodiments usingthe IBR Channel MUX 628 of FIG. 23 may not be able to resolve multiplestreams during the training Block 0 of FIG. 26 because all streamsarrive at all antennas by substantially similar paths which makescancellation for multiple demodulation streams impractical. Onealternative for the RRC 660 in this situation is to allocate one streamper link only. However, this results in reduced throughput for the linkwhich is not only undesirable but highly counterintuitive for manybackhaul operations personnel likely to perceive that unobstructed LOSshould be “better” than obstructed LOS. Thus, the RRC 660 may alsoevaluate certain antenna selection options to intentionally create pathdiversity in otherwise unobstructed or nearly unobstructed LOSconditions.

A first alternative for the RRC 660 to provide multiple streams withboth obstructed and non-obstructed LOS operation is the dynamic testingand possibly selection of mapping different modulator streams todifferent antenna elements (via separate RF chains) based on differentantenna polarizations. Because of the typically substantial signalimpairment associated with a link that is transmitting receiving fromopposite polarization antenna elements, testing of alternativepolarization antenna elements with training data may be pre-arranged intime by RRC control frames exchanged by both IBRs with an instant link.Similarly, the AE-IBR may select any changes in link antenna elementsinvolving polarization and verify an agreed upon changeover time by RRCcontrol frame exchange for reasons analogous to those for RF carrierfrequency or channel bandwidth changes. A significant advantage of usingpolarization diversity amongst the set of selectable antenna elements isthat multiple stream throughput can be maintained using a common set ofchannel equalization techniques as described above for MIMO operationwith the exemplary IBR Channel MUX of FIG. 23.

A second alternative for the RRC 660 to provide multiple streams withboth obstructed and non-obstructed LSO operation is the dynamic testingand possibly selection of mapping different modulator streams todifferent antenna elements (via separate RF chains) based on differentdirection orientations of antenna elements as is possible with exemplaryantenna arrays such as those depicted in FIGS. 14 and 15. The RRC 660 insuch IBR embodiments can test and/or change-over such multi-directionalantenna element combinations entirely at one end of a link without thenecessity of exchanging RRC control frames or requiring a coordinatedchange-over at both IBRs in a link simultaneously. For at least thepreceding reason, this second alternative to maintaining multi-streamoperation by using multi-directional antenna element combinations thatintentionally create propagation path diversity may be desirable for anRE-IBR used in a PMP configuration. An example antenna array suitablefor such an RE-IBR is depicted in FIG. 14. Furthermore, this directionalorientation diversity strategy to multi-stream operation in otherwiseunobstructed LOS is also advantageous compared to the polarizationdiversity option in PMP deployments because it does not require that theAE-IBR deploy certain resources (e.g., dedicated RF chains and antennaelements) to link modalities of benefit to some RE-IBRs but not others(that do experience obstructed LOS) as may occur with polarizationdiversity in a PMP deployment.

With reference to FIGS. 7, 32 and 33, the Intelligent BackhaulManagement System (IBMS) Agent 700 is an optional element of the IBRthat optimizes performance of the instant links at the IBR as well aspotentially other IBR links in the nearby geographic proximity includingpotential future links for IBRs yet to be deployed. As described abovein reference to the RRC 660 and depicted in FIG. 32, the primaryinteraction of the IBMS Agent 700 with the internal elements of the IBRis via a direct bi-directional path to the RRC 660. In one direction,the various policies and configuration parameters used by the RRC 660 toallocate resources within and amongst IBRs with active links to eachother are sent from the IBMS Agent 700 to the RRC 660. In the returndirection, the RRC 660 reports operational statistics and parametersback to the IBMS Agent 700 both from normal operation modes and from“probe in space” modes as directed by the IBMS Agent 700.

In contrast with the RRC 660, which communicates with other elements ofthe IBR internally or with other RRC entities at IBRs actively linked toits IBR, the IBMS Agent 700 can receive information from or transmitinformation to or initiate sessions with other elements of the overallIBMS that are logically located anywhere within any network (subject toappropriate access privileges). As shown in FIG. 32, the IBMS Agent 700appears to the overall network as an Applications layer entity thatexchanges messages typically over TCP/IP and any of the link layerinterfaces within the IBR. For the common deployment of IBRs within acellular system Radio Access Network (RAN) to backhaul cellular basestation sites, it may be necessary for the IBR to tunnel TCP/IP across acellular RAN specific transport and network layers protocol, such asGPRS Tunneling Protocol (GTP) to reach a gateway that bridges to TCP/IPand on to the desired other IBMS elements.

In some embodiments, the IBMS Agent 700 can act as an autonomous entitythat per configuration settings draws information from network resources(whether private or public) and serves as an expert system local to be aspecific IBR to optimize its performance. In other embodiments, the IBMSAgent 700 interacts with other peer IBMS Agents at other IBRs within its“interference” zone of influence via self-discovery within the immediatenetwork so that such peer IBMS Agents can collectively optimize IBRperformance within the zone. In some embodiments, the IBMS Agent 700 isa client of one or more IBMS servers that may be within the privateand/or public networks such that communications with a particular IBMSAgent 700 is always to or from or via such IBMS servers.

The information gathered at the IBR and distilled by the IBMS Agent 700regarding, for example, operational statistics (such as RSSI, channelequalization metrics, FCS failures, etc.) and resource selections (suchas antennas, channel bandwidth, modulator stream assignments, etc.), maybe sent to an IBMS server. Such information can be used by the IBMSserver to improve performance predictability of future IBR deploymentsor to enable overall IBR system performance of all links within thepurview of the IBMS server by policy optimization across IBRs. Thecommunications from the IBMS server to the IBMS Agents can include suchoptimized policy parameters based on information from other IBMS Agentsand private and/or public databases such as directories of known non-IBRlinks including their locations, antenna patterns, power levels, carrierfrequencies and channel bandwidths, tower heights, etc.

With reference to FIGS. 31 and 32, other exemplary applications layerprotocols are shown for the IBR. A Hyper Test Transfer Protocol (HTTP)server application can enable any browser (or browser-like application)on a networked computer, including a WiFi connected mobile device suchas laptop, table or smart-phone, to query or modify various IBRconfiguration parameters. The Simple Network Management System client ofFIGS. 31 and 32 is an example of an industry-standard network managementutility that can also be used to query or modify IBR configurationparameters using tools such as HP Open View.

The foregoing description of the various elements of the IBR inreference to FIGS. 5-36 have described numerous exemplary embodiments.These exemplary element embodiments may be assembled together in manydifferent combinations and permutations. A very short description of buta few of the possible overall IBR exemplary embodiments is summarizedbriefly below.

A first exemplary embodiment of an IBR includes the features outlined inTable 1.

TABLE 1 First Exemplary Intelligent Backhaul Radio Features TDDoperation PTP configuration only (AE-IBR or RE-IBR) SC-FDE modulationJmod = Jdem = 1 K = 2, M = 4 L = 2, N = 4 Q = 8, antenna array similarto FIG. 14 front facet with 2 Vertical and 2 Horizontal polarizationantenna elements at 15 dBi side facets each with 1 Vertical and 1Horizontal polarization antenna elements at 12 dBi QPSK, 16 QAM, 64 QAM,256 QAM 1/3, 1/2, 2/3, 3/4, 5/6, 7/8 coding rates 1 or 2 modulatedstreams MMSE combining of up to 4 receive chains MRC weighting of up to4 transmit chains based on blended weights per chain across entirechannel as derived from Rx conditions fixed superframe (1 ms) NACKprotocol up to 28 MHz channel bandwidth single MPDU per PPDU

This first exemplary embodiment can have a very high MAC efficiency(ratio of MPDU payload bits to overall MAC bits) under heavy loadingfrom the IBR LLC—in excess of 95%. Furthermore, the PHY efficiency(ratio of time where PPDU payload symbols excluding PAD are actuallytransmitted to superframe time) can exceed 90% for typical channelimpairment conditions and ranges of 2 km or less. At 28 MHz symbol rateand channel bandwidth, 256 QAM, 2 modulated streams, ⅞ rate coding, andwith MAC and PHY efficiencies of 95% and 90% respectively, the aggregatebi-directional throughput for this first exemplary IBR embodiment canexceed 330 Mb/s with an average end to end latency of about 2 ms.

A second exemplary embodiment of an IBR includes the features outlinedin Table 2.

TABLE 2 Second Exemplary Intelligent Backhaul Radio Features TDDoperation PMP configuration as the AE-IBR OFDM modulation Jmod = 2, Jdem= 3 K = 4, M = 4 L = 5, N = 5 Q = 9, antenna array similar to a 180°azimuth coverage “half-array” of FIG. 15 per each of 4 facets: 1Vertical and 1 Horizontal polarization antenna elements at 15 dBi on“top”: 1 non-polarized, 180° azimuth (aligned with array) coverage 8 dBielement connected only to the 5^(th) Rx chain and 3^(rd) demodulatorcore QPSK, 16 QAM, 64 QAM, 256 QAM 1/3, 1/2, 2/3, 3/4, 5/6, 7/8 codingrates up to 2 modulated streams per modulator core MMSE combining of upto 5 receive chains Eigenbeamforming transmit SDMA of up to 4 transmitchains fixed superframe (0.5 ms or 1 ms) NACK protocol TDMA alternatesuperframe per RE-IBR up to 2 RE-IBRs per modulator core up to 28 MHzchannel bandwidth single MPDU per PPDU

This second exemplary embodiment uses OFDM rather than SC-FDE in orderto enable transmit SDMA of up to 4 RE-IBRs (only 2 simultaneously) in ahighly frequency selective channel. As discussed above, SC-FDE couldalso be used in this PMP AE-IBR with theoretically similar performancebut with more complex baseband processing required. This secondexemplary embodiment should have similar MAC efficiency to the first forthe 1 ms superframe case but the PHY efficiency (which needs adefinition that accounts for data block pilot subchannels andzero-padded subchannels) is typically lower with 85% being excellent.The eigenbeamforming may also require additional overheads in manypropagation environments. If 4 RE-IBRs are used in TDMA mode, thelatency would expand to 4-5 ms on average for 1 ms superframes and abouthalf that for 0.5 ms superframes. With 2 RE-IBRs that are spatiallyseparable and with operating parameters set as described for the firstexemplary embodiment above, aggregate bi-directional throughput for thissecond exemplary embodiment can be as high as about 600 Mb/s. Note thatthe “top” antenna which is not connected to a transmit chain can be usedto provide the MMSE combiner with an additional degree of freedom tocancel interference when 4 modulated streams from 2 RE-IBRs are beingsimultaneously received. It can also be used advantageously as a “probein space” to provide information to the IBMS or to assist in DFS byscanning channels not currently used at the AE-IBR. Also note thatalthough this second exemplary embodiment can be used as an RE-IBR foritself that preferably this role may be filled by the third exemplaryembodiment described below.

A third exemplary embodiment of an IBR includes the features outlined inTable 3.

TABLE 3 Third Exemplary Intelligent Backhaul Radio Features TDDoperation PTP configuration only (AE-IBR or RE-IBR) or PMP configurationas the RE-IBR OFDM modulation Jmod = Jdem = 1 K = 2, M = 4 L = 2, N = 4Q = 8, antenna array similar to FIG. 14 front facet with 2 Vertical and2 Horizontal polarization antenna elements at 15 dBi side facets eachwith 1 Vertical and 1 Horizontal polarization antenna elements at 12 dBiQPSK, 16 QAM, 64 QAM, 256 QAM 1/3, 1/2, 2/3, 3/4, 5/6, 7/8 coding rates1 or 2 modulated streams MMSE combining of up to 4 receive chains MRCweighting of up to 4 transmit chains based on blended weights per chainacross entire channel as derived from Rx conditions fixed superframe (1ms) NACK protocol up to 28 MHz channel bandwidth single MPDU per PPDU

Note that while this third embodiment can also be used as a PTP AE-IBRand RE-IBR combination, it is unlikely to provide meaningful performanceimprovements for PTP compared to the first exemplary embodiment andquite possibly would have slightly lower aggregate bi-directionalthroughput and would require a less efficient and more expensive poweramplifier and stricter phase noise considerations. Forcommercially-available components today, using OFDM versus SC-FDE at RFcarrier frequencies above 10 GHz is extremely challenging. Note that forbelow 10 GHz operation, it is commercially feasible today to use SDRbaseband approaches and common chain and Front-end components to buildan IBR software programmable as either the PTP first exemplaryembodiment or the PMP RE-IBR third exemplary embodiment.

A fourth exemplary embodiment of an IBR includes the features outlinedin Table 4.

TABLE 4 Fourth Exemplary Intelligent Backhaul Radio Features FDDoperation PMP configuration as the AE-IBR OFDM modulation Jmod = 4, Jdem= 6 K = 8, M = 8 L = 10, N = 10 Q = 18, antenna array similar to FIG. 15per each of 8 facets: 1 Vertical and 1 Horizontal polarization antennaelements at 15 dBi on “top”: 2 non-polarized, opposite facing, 180°azimuth coverage 8 dBi elements connected only to the respective ones ofthe 9^(th) and 10^(th) Rx chains and the 5^(th) and 6^(th) demodulatorcores QPSK, 16 QAM, 64 QAM, 256 QAM 1/3, 1/2, 2/3, 3/4, 5/6, 7/8 codingrates up to 2 modulated streams per modulator core MMSE combining of upto 10 receive chains Closed-loop eigenbeamforming transmit SDMA of up to8 transmit chains fixed superframe (.5 ms or 1 ms) NACK protocol TDMAround-robin superframe order per RE-IBR up to 4 RE-IBRs per modulatorcore up to 28 MHz channel bandwidth single MPDU per PPDU

This fourth exemplary embodiment is similar to the second exemplaryembodiment except that it utilizes a larger 360° azimuth antenna arrayand FDD operation as well as 4 time slot per modulator TDMA to supportup to 16 RE-IBRs with an aggregate bi-directional throughput of about 2Gb/s after the increased overhead in efficiencies of the system areaccounted for. Latency also increases proportionately if 4 TDMA slotsare used.

A fifth exemplary embodiment of an IBR includes the features outlined inTable 5.

TABLE 5 Fifth Exemplary Intelligent Backhaul Radio Features FDDoperation PMP configuration as the RE-IBR OFDM modulation Jmod = Jdem =1 K = 2, M = 4 L = 2, N = 4 Q = 8, antenna array similar to FIG. 14front facet with 2 Vertical and 2 Horizontal polarization antennaelements at 15 dBi side facets each with 1 Vertical and 1 Horizontalpolarization antenna elements at 12 dBi QPSK, 16 QAM, 64 QAM, 256 QAM1/3, 1/2, 2/3, 3/4, 5/6, 7/8 coding rates 1 or 2 modulated streams MMSEcombining of up to 4 receive chains Selected unweighted Tx antennas asdirected by receiver at opposite end fixed superframe (1 ms) NACKprotocol up to 28 MHz channel bandwidth single MPDU per PPDUThe primary application of this fifth exemplary embodiment is to serveas an RE-IBR for the fourth exemplary embodiment AE-IBR.

A sixth exemplary embodiment of an IBR includes the features outlined inTable 6.

TABLE 6 Sixth Exemplary Intelligent Backhaul Radio Features FDDoperation PTP configuration only (AE-IBR or RE-IBR) SC-FDE modulationJmod = Jdem = 1 K = M = 2 L = 2, N = 4 Q = 6, antenna array similar toFIG. 13 3 Vertical polarization antenna elements, 2 at 13 dBi and 1 at18 dBi, 3 Horizontal polarization antenna elements, 2 at 13 dBi, 1 at 18dBi QPSK, 16 QAM, 67 QAM, 256 QAM, 1024 QAM 1/3, 1/2, 2/3, 3/4, 5/6,7/8, 9/10 coding rates 1 or 2 modulated streams MMSE combining of up to4 receive chains selection of 2 antennas in transmit fixed superframe (1ms) NACK protocol up to 56 MHz channel bandwidth single MPDU per PPDU

This sixth exemplary embodiment provides high aggregate bi-directionalthroughput of up to about 1.8 Gb/s with moderate complexity relative toother IBRs. This sixth exemplary embodiment performs optimally inpropagation channels with only moderate obstructions compared tounobstructed LOS. FDD also provides <1 ms average latency.

A seventh exemplary embodiment of an IBR includes the features outlinedin Table 7.

TABLE 7 Seventh Exemplary Intelligent Backhaul Radio Features FDD/TDDhybrid operation PTP configuration only (AE-IBR or RE-IBR) SC-FDEmodulation Jmod = Jdem = 1 K = 4, M = 8 L = 4, N = 8 Q = 12, antennaarray similar to FIG. 14 per each of 3 facets: 2 Vertical and 2Horizontal antenna elements of 15 dBi each QPSK, 16 QAM, 64 QAM, 256 QAM1/3, 1/2, 2/3, 3/4, 5/6, 7/8 coding rates up to 4 modulated streams MMSEcombining of up to 8 receive chains MRC weighting of up to 8 transmitchains based on blended weights per chain across entire channel asderived from Rx conditions fixed superframe (1 ms) NACK protocol up to56 MHz channel bandwidth single MPDU per PPDU

This seventh exemplary embodiment has additional resources compared tothe sixth exemplary embodiment to provide higher aggregatebi-directional throughput of about 3 Gb/s for a PTP link with <1 mslatency that can operate in a severely obstructed LOS propagationchannel. It advantageously uses hybrid FDD/TDD operation wherein eachfrequency duplexed channel alternates in opposite synchronization toeach other between transmit and receive. This enables a relativelystraightforward and efficient transmit chain weighting to be derivedfrom receive chain equalization analysis without increasing latency.Furthermore, an additional degree of frequency diversity (and space tothe extent different antennas are selected) is achieved. The FDD/TDDhybrid can be utilized on any FDD IBR deployment where spectrumregulations permit it. To the extent each FDD band operation relies onband-specific band-select filters in the Front-ends, then additionalcircuit complexity for switching between transmit and receive is needed.

Note that the preceding embodiments are a small subset of the possibleIBR embodiments that are enabled by the disclosure herein. Note furtherthat many additional optional structures or methods described herein canbe substituted within the above exemplary embodiments or otherembodiments.

For example, TDD CSMA could be used advantageously in high interferencespectrum allocations or where required by spectrum regulations as asubstitute for fixed superframe timing in the above exemplaryembodiments.

Note also that all of the above exemplary embodiments are compatiblewith any RF carrier frequencies in the range of interest fromapproximately 500 MHz to 100 GHz.

Note further that in multi-channel embodiments, it is possible to usedifferent access and MAC protocols in different channels especiallywhere advantageous for or required by spectrum regulations. For example,an IBR link may advantageously provide a “base” throughput capability ina channel expected to have minimal interference but for which licensingcosts or regulatory restrictions limit total throughput. Then a “surge”throughput capability can be provided in a second channel, such asunlicensed spectrum, where throughput can be higher but the risk oftemporal interference outages is also higher.

As evident from the above exemplary embodiments, OFDM is typically usedfor PMP deployments because at a baseband processing level it isrelatively less complex than SC-FDE if frequency-selective channels areto be used with transmit SDMA (at least at the AE-IBR). However, OFDMhas higher peak to average ratio and is more sensitive to carrierfrequency offset and phase noise than SC-FDE. Thus, for PTP, SC-FDE isoften preferable, especially for operation at RF carrier frequenciesabove 10 GHz where commercially viable components are expensive foreither OFDM power amplification or OFDM-compatible local oscillatorspecifications.

Note that in backhaul applications where links are nominally continuous,additional techniques can improve PHY efficiency. For example, withOFDM, the training block can be combined with a PLCP Header block byinterleaving subchannels appropriately. This is also possible for SC-FDEif DFT pre-coding (i.e. Tx block Assembler-k includes a DFT and Tx-Mux-mincludes an IDFT) is used. DFT pre-coding for SC-FDE can also be used topulse shape the transmitted waveform similar to OFDM zero padding and/orwindowing. The training block in SC-FDE can also be shorter than thedata blocks to save PHY overhead by either using another FFT fortraining or switching the FFT bin size during the training block. Thechannel equalization function so derived is then interpolated for use onthe longer data blocks with additional frequency bins. Also, when usingDFT pre-coding in SC-FDE, it is possible to time multiplex an FFT blockbetween transmit and receive or within transmit or receive such that allfour FFT operations of an SC-FDE transceiver can be realized by 2 oreven 1 FFT hardware core. Another technique to simplify PMP deploymentof IBRs is to use OFDM in the forward link from AE-IBR to the multitudeof RE-IBRs, and then use SC-FDE in the reverse link from RE-IBR toAE-IBR. This enables the advantages of transmit eigenbeamforming at theAE-IBR in a frequency-selective channel based primarily on receiveequalization while keeping the RE-IBRs relatively straightforward withmuch simpler transmitters than in the OFDM only case.

Numerous additional variations of the above-described elements of theIBR can also be advantageously utilized in substitution for or incombination with the exemplary embodiments described above. For example,antenna elements need not always be shared between transmit and receivewhether in TDD or FDD mode. In certain embodiments, it is preferable tohave a smaller number of transmit antenna elements, often with broaderazimuthal coverage than those used by the receive antenna elements, thatare always used versus a selectable larger number of receive antennaelements. In some embodiments with separate transmit and receive antennaelements, the respective front-ends of FIGS. 11 and 12 would not requirerespectively the SPDT switch or the band selection within a duplexerfilter and either the receive path elements for a front-end coupled to atransmit antenna element or the transmit path elements for a front-endcoupled to a receive antenna element, which further advantageouslyreduces cost and complexity of the IBR. In such embodiments, eachtransmit antenna element is coupled to a respective transmit poweramplifier which is in turn coupled to a respective transmit RF chainwherein such couplings may be selectable RF connections or fixed RFconnections—the latter for IBR embodiments wherein certain transmit RFchains are always connected to certain transmit power amplifiers andcertain transmit antenna elements whenever the IBR is in a transmitmode. Similarly, each receive antenna element is coupled to a respectivereceive low noise amplifier which is in turn coupled to a respectivereceive RF chain wherein such couplings are typically selectable RFconnections as described herein.

As another example, the NACK protocol described above with reference toFIG. 33 and various IBR embodiments can be extended to either ACK/NACK,single NACK as described above, or persistent NACK until delivered.Furthermore, such exemplary choices can be applied to individual MSDUsbased on either an indicator from the IBR LLC of FIG. 33 or inspectionof MSDU class of service or type of service header field bits as definedby a policy either within the IBR MAC or updateable via the optionalIBMS Agent 700. For example, MSDUs corresponding to Voice over InternetProtocol (VoIP) packets are typically sent at a high class of servicepriority but with so little tolerance for latency that it may bepreferable to send such MSDUs with no ACK/NACK retransmission option orwith only the single NACK protocol described previously. At the oppositeextreme, certain data transfers, such as for example only, cellularnetwork control or management plane messages or user file transfer data,may tolerate considerable or unpredictable latencies associated withpersistent retransmission until NACK=0 rather than rely on much slowerupper layer retransmission protocols. The policy for mapping MSDUs to aparticular ACK/NACK strategy may also be responsive to radio channelconditions and/or loading as determined within the various elements ofthe IBR described above. For example, when the current packet failurerate is very low and/or the loading demand on the IBR is high comparedto the capacity of the MCS, then one policy may be to minimizeretransmissions by transmitting most MSDUs with no ACK/NACK or a singleNACK. Alternatively, for a high packet failure situation and/or lowdemand at a given MCS, the opposite strategy may be used. Also, any ofthe IBR embodiments described herein that use copper Ethernet interfacesmay also use such interfaces to supply Power over Ethernet (PoE) to theIBR.

One exemplary embodiment of an antenna system for the IBR is based on acombination of angle-beam diversity and polarization diversity. In someembodiments, an advantageous far-field radiation pattern may be achievedwith multiple instances of a two-port, dual-polarity sector antennaarranged side-by-side at various angles.

An exemplary antenna is illustrated in FIG. 37. The correspondingoverlaid far-field radiation patterns for the antenna of FIG. 37 areillustrated in FIG. 38. It should be noted that the far-field patternsare identical for both polarizations, and for convenience are shownutilizing a single plot for both polarizations. It should be noted thatsuch convention is utilized where applicable in subsequent diagrams andplots as well.

In FIG. 37, each antenna is used as bi-directional (i.e., for bothtransmit and receive). Each antenna may be used in atime-division-duplex system or frequency division duplexed system.Signals 3710 (A,B,C,D) are respectively transmitted and received by twoport, dual polarized sector antennas 3701 (A,B,C,D), which are coupledto base band signal processing 3730 respectively by transceivers 3720(A,B,C,D). It will be appreciated that the antenna system of FIG. 37 maybe employ any and all techniques disclosed herein.

FIG. 38 is a plot of an exemplary far field antenna pattern of a sectorantenna panel. Patterns 3801(A,B,C,D) respectively correspond to sectorantennas 3701(A,B,C,D). The bore sight of the antenna system(corresponding to zero degrees alignment) of FIG. 37 corresponds to boresight 3820 in FIG. 38, with each incremental radial referencecorresponding to 30 degree increments. In FIG. 38, increasing referencegain increments 3815, 3810, 3805 correspond to steps of 5 dB.

FIG. 39 illustrates an alternative sector antenna panel and processing,including separate transmit and receive antennas. The arrangement ofFIG. 39 includes dedicated unidirectional transmit/receive antennas,which allows the patterns to be set independently. Dedicatedunidirectional transmit/receive antennas can be advantageous in certaintypes of regulated bands, such as EIRP limited bands. The arrangementshown in FIG. 39 is similar to the arrangement of FIG. 37, but anadditional two-port, dual polarity transmit antenna 3950 is provided asa forward-facing center panel. In FIG. 39, the four other two-port dualpolarity antennas 3901 (A,B,C,D) are used as receive-only antennas.

FIG. 40 is a plot of an exemplary far field antenna pattern of thealternative sector antenna panel including separate transmit and receiveantennas of FIG. 39. The transmit sector panel 3950 has a widerazimuthal beam width so that the transmit beam width 4030 issubstantially equal to the composite receive antenna beam width4001(A,B,C,D).

FIG. 41 illustrates an alternative sector antenna panel and processingincluding separate transmit and receive antennas. The antenna panel ofFIG. 41 has an alternative geometric arrangement of elements. In FIG.41, the order of the receive antennas 3901(A,B,C,D) in the antenna arrayis re-assigned. In FIG. 39, the inner-most receive antennas (3901B andC) are arranged to the outside of the array (4101A,D) and increased inforward angular orientation such that the overlap of the antennapatterns relating to antennas (4101A,D) is increased. In such anarrangement, the antennas 4101A,B,C,D correspond respectively to theantenna patterns of FIG. 42 is a differing order of antenna 4201B,A,D,Cdue to the aforementioned modifications in arrangement and orientation.Spatial diversity is achieved due to the additional physical distancebetween antennas 4101A and 4101D despite the overlap of the antennapatterns 4201 B and 4201C. The transmit antenna pattern of antenna 4150is identical in FIG. 42 to the pattern of antenna 3950 as shown in plot4030.

FIG. 43 illustrates a dual-polarity, two-port patch antenna elementincluding feed and grounding points. The arrangement of FIG. 43 may beused as a component of the antenna arrays described herein for use as areceive or a transmit antenna panel. In FIG. 43, the dual-port design isbased on a circular patch antenna 4300. In FIG. 43, two orthogonal modesare excited by two orthogonal probe feeds 4320 and 4310. Each modeexcites linearly polarized far-field radiation. A shorting pin 4330 isprovided in the center of the patch to suppress the DC-mode of the patchthat would normally be the primary mechanism creating undesired couplingbetween the two ports. The construction is based on two microwave-gradePCBs: one is used as a ground-plane 4302 for the patch, and the othercontains the etched circular patch 4301. The ground-plane PCB 4302 alsoprovides micro strip feed structure to feeds 4320 and 4310.

FIG. 44A and FIG. 44B illustrate an exemplary dual-polarity, two port,patch antenna array. The antenna array includes etched patch antennaelements 4401A-H. In FIGS. 44A and 44B, the antenna elements 4401A-Hinclude patch 4301. A shared patch array ground plane 4402, whichcorresponds to the ground plane 4302, is provided for each patchelement. Each patch element 4401A-H includes a two port antenna elementutilizing orthogonal polarization modes. In one embodiment, feeds4410A-H are provided for a first polarization, and feeds 4420A-H areprovided for the second polarization.

In some embodiments, the collective patch antenna array 4400A providesfor a collective two port interface and provides for a common microstrip cooperate feed network integrated with the ground plane 4403 PCB.For example, a first port feeds each of the same polarization feeds4410A-H providing for a polarized sub-array of array 4400A, while asecond port feeds each of the other same polarization feeds 4420A-Hproviding for the other polarized sub-array. In some embodiments, amicro strip cooperative feed network is provided for each polarizationto have a common delay from each array port to each of the respectivepolarization feeds to achieve a desired array factor, and, in turn, adesired array far-field pattern. In some embodiments, variations on therelative array port to polarization feeds are provided to achievevarious modified antenna patterns such as a modified beam width, sidelobe levels, or the like.

In some embodiments, the antenna array also includes a grounded fencestructure 4440A and 4440B which provides advantages relating to improvedazimuthal array gain directivity, and side lobe levels, as well aspotentially improved near field isolation from other additional antennastructures. In some embodiments, the antenna array includes supportingstructures 4405A-J to provide structural support of printed patch PCB4402 from ground plane 4403 PCB. It will be appreciated that the patcharray 4400A may be used in the receive or transmit panels of FIG. 41, aswell as other embodiments disclosed herein.

FIG. 45A and FIG. 45B illustrate an exemplary single-polarity, singleport, printed dipole antenna element 4500. The antenna element 4500 maybe used in an antenna array. FIG. 45A and FIG. 45B show more than asingle representative element; however, only a single element 4500 isdescribed with reference to FIGS. 45A and 45B.

In FIGS. 45A and 45B, the dipole antenna element 4500 is etched onprinted circuit board 4502. A first arm of the element 4510 is connectedto a micro strip feed network integrated with PBC 4502 via a BALUNstructure element 4514. The BALUN structure includes structures 4514 and4507. Structures 4514 and 4507 receive a unbalanced signal from themicro strip feed network 4516 utilizing post 4515. The associated post4515 couples a signal to the BALUN element 4514, which produces abalanced signal (together with BALUN element 4507), which then iscoupled to balanced impedance matching structures and 4511 and 4506. Thepost passes through ground plane 4501 via a clearance structure 4503. Anelement arm 4505 acts as a counter poise to element arm 4510, andconnects to BALUN structure 4507 via impedance matching 4506. A groundplane 4501 is provided to provide additional directive gain in the Zaxis. Collectively, the printed dipole element 4500 provides a singleport, single linear polarization directive antenna radiation pattern. Insome embodiments, the BALUN ground plane 4507 is electrically connectedto the reflector ground plane 4501 to ensure a low-impedance transitionfrom the backplane microstrip structure to the antenna microstripstructure. In the embodiment shown in FIGS. 45A and 45B, the arms of theprinted dipole are swept back primarily to widen the beamwidth of eachelement. A secondary benefit of swept back arms is that the impedancebandwidth (VSWR bandwidth) of the antenna is improved.

FIG. 46A and FIG. 46B illustrate an exemplary dual-polarity, two port,antenna array utilizing printed dipole antenna elements. In someembodiments, the antenna elements 4605A-J are equivalent to theexemplary single-polarity, single port, printed dipole antenna element4500 shown in FIG. 45A and FIG. 45B. Elements 4605A,C,E,G, and I arerotated at a positive 45 degrees, providing for a positive “slant 45”polarization “sub-array.” The elements 4605A,C,E,G, and I are commonlyfed with a shared cooperate feed network and are coupled to a first portof array 4600A. Elements 4605B,D,F,H, and J are rotated at a negative 45degrees, providing a negative “slant 45” polarization sub-array. Theelements 4605B,D,F,H, and J provide for the opposite orthogonallypolarized sub-array. Elements 4605B,D,F,H, and J are commonly fed withanother shared cooperate feed network and are coupled to the othersecond port of array 4600A. In some embodiments, the positive 45 degreeslant dipoles are placed offset form the negative 45 degree slantdipoles to minimize the mutual coupling from any of elements4605B,D,F,H, and J to any nearby elements in the set 4605A,C,E,G, and I.Additionally, the placement of the elements 4605B,D,F,H, and J are thenulls of elements 4605A,C,E,G, and I and vice versa. Such an arrangementis achieved by placing the center of each element of one column alignedat 90 degrees to dipole arms of the opposite column, and havingappropriate offset spacing so as to not couple the elements.

FIG. 46C illustrates the printed circuit board providing for a groundplane of an exemplary dual-polarity, two port, antenna array utilizingprinted dipole antenna elements, showing a cooperate feed network foreach single polarized sub-array associated with each port. The antennaarray includes a first array port 4640 to couple a signal for a firstpolarized sub-array utilizing micro-strip lines and divider structures.The antenna array includes a second port 4620 to couple a signal for asecond orthogonal polarization utilizing micro-strip lines and dividerstructures.

In some embodiments, the group delay from port 4640 to each associatedantenna element feed posts 4615B,D,F,H,J is equalized. In someembodiments, each feed network has equalized electrical lengths suchthat each port is excited in-phase. Likewise, in some embodiments, thegroup delay from port 4620 to each associated antenna element feed posts4615A,C,E,G,I is designed to also be equalized. In some embodiments, thesignal power coupled to or from each of the array ports 4640, 4620 witheach of the associated antenna sub-array elements is equal. Suchsplitting or combining and delay equalization is performed utilizingmicro-strip dividers and impedance matching structures, such as 4660,4655, 4652, 4645, 4650, 4625, and 4630. In some embodiments, the delaysof each of the feed networks are equal to each other, or have apre-determined known different. Such embodiments provide for a reducedcomplexity when performing specific transmission or reception beamforming operations, or adaptive polarization operations. Additionally,such offsets may be compensated for in digital baseband, or in other RFdelay equalization components. One such structure for adding delay orphase offsets is the transmit channel equalizer Tx-CE-m 2308.Additionally, in some embodiments, each polarizer sub-array (fed byports 4640 and 4620 respectively) is fed with separate transmitterchains utilizing different transmit channel equalizers 2308 within theIBR Channel MUX 628 as shown in FIG. 23A.

In some embodiments, adjustable polarization is provided by defining therelationship of the respective Tx-CE 2308 weights {right arrow over(WT)}_(m,k) feeding each of the polarized sub-arrays of the same arraypanel as applied to the transmit symbol stream {right arrow over(TxBlock)}_(k). For example, transmitting a Tx Block solely from onepolarized sub-array relative to the other results in the respectivepolarization alone. However, transmitting a Tx Block with the same phasefrom each of the cross polarized sub-arrays results in a polarizationwhich is linear and at 45 degrees between the two respectivepolarizations. Further adjustment of the “slant” of the resultingpolarization may be achieved utilizing various relative amplitudesbetween the two equalized transmit blocks.

Additional TX polarizations may be achieved by adjusting the relativeamplitude and phase between the same transmit block {right arrow over(TxBlock)}_(k) on each of the cross polarized sub-arrays of the samepanel. As an additional example, circular polarized transmission may beachieved by delaying the phase of one transmitted signal by 90 degreesfor right handed polarization (or −90 degrees for left handed circular).Such adjustments may be made adaptively based upon various criteria,including metrics feedback from the target receiving IBR. Suchprocessing may be extended to antenna arrays and sub-arrays which arenot within the same array or panel. Such processing results intransmitter beam forming operations. Further, each transmit streams ortransmit block {right arrow over (TxBlock)}_(k) may be adjustedindependently as described in relation to embodiments of FIG. 24 andFIG. 24A, and may provide for frequency selective block specificadaptive or adjustable polarization transmission or other forms of beamforming.

FIG. 47A illustrates a plot of an exemplary far field elevation antennapattern of a dual-polarity, two port, antenna array utilizing printeddipole antenna elements, and FIG. 47B illustrates a plot of an exemplaryfar field azimuthal antenna pattern of a dual-polarity, two port,antenna array utilizing printed dipole antenna elements. In someembodiments, the antenna patterns for each of the polarized sub-arraysprovide substantially the same pattern; accordingly, only onerepresentative plot is shown representing both sub-arrays.

FIG. 48 illustrates an alternative exemplary dual-polarity, two port,antenna array utilizing printed dipole antenna elements, includingalternative printed dipole antenna elements and further providingvertical and horizontal polarizations. Array 4800 provides forvertically and horizontally polarization, rather than the slantpolarizations of 4600A. The arrangement of the elements in thealternative array configurations of 4800 are modified to minimize themutual coupling from any of the elements 4810A,B,C,D, and E to any ofthe elements are within the antenna pattern nulls of elements 4820A,B,C,and D. There is strong mutual coupling between the elements in eachsubset (4810A,B,C, D, and E) and (4820 A, B, C, and D), but thiscoupling is not detrimental to the operation of a uniformly excited(both amplitude and phase) antenna array, but the opposite is not truein this arrangement.

Elements 4820A,B,C,D are within the antenna pattern of the elements ofthe orthogonally polarized sub array 4810A,B,C,D and E. However, becauseelements 4810A,B,C,D and E are oriented at 90 degrees to the field linesof the elements 4820A,B,C,D, they do not substantially interactelectromagnetically. The parasitic elements 4905A and B increase theazimuthal beamwidth of the horizontal polarized antenna element beyondthat achievable with a conventional dipole. The parasitic elements 4905Aand B also improve the impedance bandwidth (VSWR bandwidth) of theantenna element, and in turn, the bandwidth of the array.

FIG. 49A illustrates an alternative exemplary printed dipole antennaelement. In some embodiments, antenna 4900A includes elements4810A,B,C,D and E. Antenna 4900A includes a printed circuit board 4901that is etched to provide two branches of a dipole antenna arms 4902 and4903, balanced feed ports 4925A, 4925B, directors 4905A, 4905B andsecuring tabs 4915A and 4915B.

In some embodiments, the length and position of the parasitic directors4905A, 4905B relative to the driven dipole is optimized to achieve adesired azimuthal radiation bandwidth. The position may be optimized byvarying the distance between the end of the dipole branches 4903, 4902and the parasitic elements 4905A, 4905B, as well as the fore and aftoffset of the parasitic element 4905A, 4905B relative to the dipolebranches 4903, 4902.

The antenna also includes tabs 4910A,B and 4920A and B, which are usedto secure the antenna element to the ground plane PCB 4801, and aresoldered in place as shown in FIG. 49B. Each tab mates with a plattedthrough hole slot in the PCB 4945, which allows for a securing solderjoint. In some embodiments, the soldering is performed with a wavesoldering operation. In some embodiments, the soldering is performedusing a solder paste reflowed, as typically used in surface mountoperations, and wicked into the associated joints. In some embodiments,adhesive or epoxy is used to form one or more of the tab PCB joints.

In some embodiments, a slot 4935 is provided to provide a mechanicalconnection with an alternative implementation of antenna elements4820A,B,C, and D, as shown in FIG. 50 and FIG. 51. Antenna elements4820A,B, C and D may be fabricated on a single printed circuit board5001. Slots 5005A-E are provided to mate with slot 4935 of antenna 4900Afor mechanical rigidity. Tabs 5015A-D and 5025A-D are secured to PCB5110, as described with respect to FIGS. 49A and B. Printed dipoleelement arms 5010A-D, and 5020A-D are etched in printed circuit board5001 to form the antenna sub-array structure 5000.

FIG. 51 illustrates an exemplary assembly a dual-polarized, two port,printed dipole antenna array utilizing a printed dipole structure.Printed circuit board 5101 receives tabs associated with sub-array board5000 and elements 4900A as described above. In some embodiments, thevertical element spacing is 0.65 of a free-space wavelength (equivalentto 35 mm). In FIG. 51, the vertical polarized elements are not sweptback to minimize elevation beam width. The vertical elements featureflared out width on the dipole branches to improve the impedancebandwidth (allowing for a similar impedance bandwidth to that of thehorizontal elements). In the elevation plane, the beam width of thevertical element pattern is narrower than the beam width of thehorizontal element pattern. This difference in element pattern beamwidth is compensated in the array factors. In some embodiments, thevertical polarization array uses four elements and the horizontalpolarization array uses five elements. The extra element in thehorizontal array helps to increase the horizontal array factor (i.e.narrow the pattern beam width).

FIG. 52A illustrates an exemplary horizontally arranged intelligentbackhaul radio antenna array, FIG. 52B illustrates an alternative viewof an exemplary horizontally arranged intelligent backhaul radio antennaarray, and FIG. 52C illustrates a top of an exemplary horizontallyarranged intelligent backhaul radio antenna array. Embodiments thearrangement of FIGS. 52A-C include the features of the antenna array ofFIG. 41.

In some embodiments, the receive arrays 5210A-D are configured in anarrangement of FIG. 41, wherein the dedicated two port dual polarizedreceive arrays 4101A-D are implemented as the exemplary dual-polarity,two port, patch antenna array 4400A as shown in FIG. 44A and FIG. 44B.The dedicated transmit array of FIG. 52A may be implemented as theexemplary dual-polarity, two port, antenna array utilizing printeddipole antenna elements 4600A of FIGS. 46A,B, and C, with the additionof tabs 5201 and slots 5205. Such tabs and slots provide for the shapingof the directive patterns of one column of polarized elements (e.g.,sub-array elements 4605A,C,E,G,I) relative to the other column (e.g.,sub-array elements 4605B,D,F,H,J). The tabs on each side of the array4600A allow for the patterns of the two offset transmitter antennacolumns to have substantially centered and equal radiation patterns. Itwill be appreciated that if no slots are utilized, the ground planewould appear larger on one side of the columns than on the other side ofa specific column.

Exciting the dipoles 4605 with an applied RF signal causes inducedsurface currents on the backplane 4600A which are oriented in the samedirection as the source dipole. The slots closest to each column areparallel with the surface currents and do not significantly impede thecurrent flow. The slots located at a further distance from the dipolesare oriented orthogonally to the surface currents, causing the groundplane to look effectively narrower on the far side. The slots and tabsallow for the oppositely aligned antenna elements to not be effected bythe “wider” ground plane of the tabs, where the “aligned” tabs willappear as ground to the aligned (the nearer) element column. Such anarrangement substantially equalizes the radiation patterns of each ofthe polarized sub-arrays. For example, sub array antenna elements4605B,D,F,H, and J of 4600A, as shown in FIG. 46A, are rotated at anegative 45 degrees, providing for a negative “slant 45” polarizationsub-array. The elements 4605B,D,F,H, and J are aligned with tab 5201 andslot 5205. The area of the tabs on the side of the ground plane isadvantageous in effecting the radiation pattern because the featuresappear as a solid ground to the sub-array elements 4605B,D,F,H, and Jand appear open-circuited to the oppositely aligned sub-array elements4605A,C,E,G,I.

The receive two port, dual polarized patch antennas arrays 5210A,B,C,and D correspond to the idealized antenna patterns of FIG. 42,respectively, as 4201B,A,D,C. The transmit pattern of the two port, dualpolarization printed dipole array of 4600A was shown in relation to FIG.47A and FIG. 47B, and is further shown with an associated coverage as4030 in FIG. 40, with respect to such a transmit pattern substantiallymatching the cumulative receive patterns of arrays 5210A,B,C, and D.Additionally, the elevation patterns of 5210A,B,C, and D substantiallymatch the elevation patterns of 4600A. As noted, each array 5210A,B,C,Dand 4600A includes two ports coupling to associated oppositely polarizedsub-arrays. The antenna array includes a radome 5220, which may be madeof any radiolucent material as known in the art.

FIGS. 53A,B, C and D illustrate an exemplary vertically arrangedintelligent backhaul radio antenna array. The antenna array of FIGS.53A-D physically shifts the receiver arrays from FIG. 52A-C so that,rather than being linearly aligned horizontally, two of the receivearrays are stacked above the transmitter array and two are stacked belowthe transmitter array in a vertical configuration. Variations of thisembodiment may include all four receive arrays being in a verticalalignment, or other combinations.

The two port dual polarized patch array 4400A is utilized with dedicatedreceive arrays 5330A,B,C, and D. In contrast to the antenna array ofFIGS. 52A-C, the transmitter array 4600A is replaced in FIGS. 53A-D withthe alternative exemplary dual-polarity, two port, antenna arrayutilizing printed dipole antenna elements, including alternative printeddipole antenna elements and further providing vertical and horizontalpolarizations 4800. Either transmit array or other variations may beutilized with any or all of the exemplary antenna array embodimentsdisclosed herein.

In some embodiments, the total number of receiver sub-arrays which areavailable for use in the reception of signal, may exceed the number ofIBR receivers present in a specific radio configuration. As discussed inrelation to FIG. 10 an IBR RF Switch Fabric 1012 may be utilized coupleselectable sub-arrays to specific receivers RF-RX-l to RF-RX-N. It willbe appreciated that any and all variations disclosed herein may be usedwith the embodiments of FIGS. 52A-C. In addition, the embodiments ofFIGS. 13-15 may be used with the embodiments of FIGS. 52A-C and theembodiments of FIGS. 53A-D.

As explained above, in some embodiments, the Switch Fabric 1012 of IBRRF 628 may be controlled by the radio resource controller (RRC). It willbe appreciated that the functions and capabilities of the RRC disclosedherein may be used with the embodiments of FIGS. 52A-C and FIGS. 53A-D.

In some embodiments, the delays of each of the two feed networks areequal to each other, or are a pre-determined known different. Suchembodiments provide for a reduced complexity when performing specifictransmission or reception beam forming operations, or adaptivepolarization operations. Additionally such offsets may be compensatedfor in digital baseband, or in other RF delay equalization components.One such structure for adding delay or phase offsets is the transmitchannel equalizer Tx-CE-m 2308. Additionally, in some embodiments, eachpolarizer sub-array (fed by ports 4640 and 4620 respectively) is fedwith separate transmitter chains utilizing different transmit channelequalizers 2308 within the IBR Channel MUX 628 as shown in FIG. 23A.

Adjustable polarization may be provided by defining the relationship ofthe respective Tx-CE (2308) weights {right arrow over (WT)}_(m,k)feeding each of the polarized sub-arrays of the same array panel asapplied to the transmit symbol stream {right arrow over (TxBlock)}_(k).For example, transmitting a Tx Block solely from one polarized sub-arrayrelative to the other results in the respective polarization alone.However, transmitting a Tx Block with the same phase from each of thecross polarized sub-arrays results in a polarization which is linear andat 45 degrees between the two respective polarizations. Furtheradjustment of the “slant” of the resulting polarization may be achievedutilizing various relative amplitudes between the two equalized transmitblocks.

Additional TX polarizations may be achieved by adjusting the relativeamplitude and phase between the same transmit block {right arrow over(TxBlock)}_(k) on each of the cross polarized sub-arrays of the samepanel. As an additional example, circular polarized transmission may beachieved by delaying the phase of one transmitted signal by 90 degreesfor right handed polarization (or −90 degrees for left handed circular).Such adjustments may be made adaptively based upon various criteria,including metrics feedback from the target receiving IBR. For example,the target IBR may monitor any number of receiver metrics, including,for example, receive signal RSSI, C/I, E_(B)/N_(o), E_(B)/I_(o), FER,BER, propagation channel state information (CSI), MIMO stream to streamchannel conditioning (ease of spatial multiplexing and channelcorrelation matrix conditioning), receive polarization or angularspectrum relative to interferer polarization or angular spectrum, or thelike. All, some, or a derived metric from a target IBR may be feedbackto the transmitting IBR, and/or derived directly for TDD or otherreciprocal channel duplexing approaches. Such metrics may be used tocalculate, derive, or adaptively determine the transmission equalizerweights of 2332 for frequency flat or frequency selective transmissionequalization approaches. Such processing may be extended to antennaarrays and sub-arrays which are not within the same array or panel. Suchprocessing may result in transmitter beam forming operations. Further,each transmit streams or transmit block {right arrow over (TxBlock)}_(k)may be adjusted independently as described in relation to embodiments ofFIG. 24 and FIG. 24A, and may provide for frequency selective blockspecific adaptive or adjustable polarization transmission or other formsof beam forming.

One or more of the methodologies or functions described herein may beembodied in a computer-readable medium on which is stored one or moresets of instructions (e.g., software). The software may reside,completely or at least partially, within memory and/or within aprocessor during execution thereof. The software may further betransmitted or received over a network.

The term “computer-readable medium” should be taken to include a singlemedium or multiple media that store the one or more sets ofinstructions. The term “computer-readable medium” shall also be taken toinclude any medium that is capable of storing, encoding or carrying aset of instructions for execution by a machine and that cause a machineto perform any one or more of the methodologies of the presentinvention. The term “computer-readable medium” shall accordingly betaken to include, but not be limited to, solid-state memories, andoptical and magnetic media.

Embodiments of the invention have been described through functionalmodules at times, which are defined by executable instructions recordedon computer readable media which cause a computer, microprocessors orchipsets to perform method steps when executed. The modules have beensegregated by function for the sake of clarity. However, it should beunderstood that the modules need not correspond to discreet blocks ofcode and the described functions can be carried out by the execution ofvarious code portions stored on various media and executed at varioustimes.

It should be understood that processes and techniques described hereinare not inherently related to any particular apparatus and may beimplemented by any suitable combination of components. Further, varioustypes of general purpose devices may be used in accordance with theteachings described herein. It may also prove advantageous to constructspecialized apparatus to perform the method steps described herein. Theinvention has been described in relation to particular examples, whichare intended in all respects to be illustrative rather than restrictive.Those skilled in the art will appreciate that many differentcombinations of hardware, software, and firmware will be suitable forpracticing the present invention. Various aspects and/or components ofthe described embodiments may be used singly or in any combination. Itis intended that the specification and examples be considered asexemplary only, with a true scope and spirit of the invention beingindicated by the claims.

What is claimed is:
 1. A backhaul radio for exchanging one or more datainterface streams with one or more other backhaul radios, said backhaulradio comprising: one or more demodulator cores, wherein eachdemodulator core is configured to demodulate one or more of a pluralityof receive symbol streams to produce one or more receive data interfacestreams; a plurality of receive radio frequency (RF) chains, whereineach receive RF chain is configured to convert from a respective one ofa plurality of receive RF signals within a receive frequency band to arespective one of a plurality of receive chain output signals; afrequency selective receive path channel multiplexer, interposed betweenthe one or more demodulator cores and at least the plurality of receiveRF chains, wherein the frequency selective receive path channelmultiplexer is configured to generate the plurality of receive symbolstreams from at least the plurality of receive chain output signals; oneor more modulator cores, wherein each modulator core is configured tomodulate one or more transmit data interface streams to produce one ormore of a plurality of transmit symbol streams; a plurality of transmitradio frequency (RF) chains, wherein each transmit RF chain isconfigured to convert from a respective one of a plurality of transmitchain input signals to a respective one of a plurality of transmit RFsignals within a transmit frequency band; a transmit path channelmultiplexer, interposed between the one or more modulator cores and atleast the plurality of transmit RF chains, wherein the transmit pathchannel multiplexer is configured to generate the plurality of transmitchain input signals from at least the plurality of transmit symbolstreams; a plurality of directional antenna arrays, wherein eachdirectional antenna array comprises a plurality of directional antennasub-arrays, and wherein each directional antenna sub-array comprises aplurality of directional antenna elements; and a plurality of duplexerfilters, wherein each duplexer filter comprises at least a receiveband-select filter configured to selectively pass RF signals within thereceive frequency band and a transmit band-select filter configured toselectively pass RF signals within the transmit frequency band, whereineach duplexer filter is couplable or coupled to at least one of theplurality of directional antenna sub-arrays, wherein the receiveband-select filter of each duplexer filter is couplable or coupled to atleast one of the plurality of receive RF chains, and wherein thetransmit band-select filter of each duplexer filter is couplable orcoupled to at least one of the plurality of transmit RF chains; whereineach of the plurality of directional antenna arrays is configured tooperate over at least both of the transmit frequency band and thereceive frequency band; and wherein each of the plurality of directionalantenna arrays is horizontally offset relative to at least one other ofthe plurality of directional antenna arrays.
 2. The backhaul radio ofclaim 1, wherein each one of the one or more demodulator cores comprisesat least a decoder and a soft decision symbol demapper; and wherein eachone of the plurality of receive RF chains comprises at least a vectordemodulator and two analog to digital converters that are configured toproduce the respective one of the plurality of receive chain outputsignals, each said respective one of the plurality of receive chainoutput signals comprised of digital baseband quadrature signals.
 3. Thebackhaul radio of claim 2, wherein each one of the one or more modulatorcores comprises at least an encoder and a symbol mapper; and whereineach one of the plurality of transmit RF chains comprises at least avector modulator and two digital to analog converters that areconfigured to produce the respective one of the plurality of transmit RFsignals, each said respective one of the plurality of transmit chaininput signals comprised of digital baseband quadrature signals.
 4. Thebackhaul radio of claim 3, wherein each one of the one or moredemodulator cores comprises at least one of a descrambler or adeinterleaver; and wherein each one of the one or more modulator corescomprises at least one of a scrambler or an interleaver.
 5. The backhaulradio of claim 1, further comprising: one or more selectable RFconnections that are configured to selectively couple certain of theplurality of directional antenna sub-arrays to either or both of certainof the plurality of receive RF chains or certain of the plurality oftransmit RF chains; wherein the number of directional antenna sub-arraysthat are configured to be selectively coupled to receive RF chainsexceeds the number of receive RF chains that are configured to acceptreceive RF signals from the one or more selectable RF connections; orwherein the number of directional antenna sub-arrays that are configuredto be selectively coupled to transmit RF chains exceeds the number oftransmit RF chains that are configured to provide transmit RF signals tothe one or more selectable RF connections.
 6. The backhaul radio ofclaim 5 wherein at least one of the one or more selectable RFconnections comprises at least one RF switch.
 7. The backhaul radio ofclaim 5, wherein the set of receive RF chains that is configured toaccept receive RF signals from the one or more selectable RF connectionsis divided between a first subset that is configured to accept receiveRF signals from directional antenna sub-arrays with a first polarizationand a second subset that is configured to accept receive RF signals fromdirectional antenna sub-arrays with a second polarization; or whereinthe set of transmit RF chains that is configured to provide transmit RFsignals to the one or more selectable RF connections is divided betweena third subset that is configured to provide transmit RF signals todirectional antenna sub-arrays with a first polarization and a fourthsubset that is configured to provide transmit RF signals to directionalantenna sub-arrays with a second polarization.
 8. The backhaul radio ofclaim 1, wherein the plurality of directional antenna arrays arearranged on a plurality of facets with one or more directional antennaarrays per facet, and wherein each facet is oriented at a differentazimuth angle relative to at least one other facet.
 9. The backhaulradio of claim 1, further comprising: a plurality of power amplifiers,wherein each power amplifier is configured to amplify at least one ofthe transmit RF signals, and wherein each power amplifier is couplableor coupled to at least one of the plurality of transmit RF chains and toat least one transmit band-select filter of the plurality of duplexerfilters; and a plurality of low noise amplifiers, wherein each low noiseamplifier is configured to amplify at least one of the receive RFsignals, and wherein each low noise amplifier is couplable or coupled toat least one of the plurality of receive RF chains and to at least onereceive band-select filter of the plurality of duplexer filters.
 10. Thebackhaul radio of claim 1, wherein both of the transmit frequency bandand the receive frequency band are within a frequency range of between 2GHz and 6 GHz.
 11. The backhaul radio of claim 1, wherein both of thetransmit frequency band and the receive frequency band are within afrequency range of above 10 GHz.
 12. The backhaul radio of claim 1,wherein the frequency selective receive path channel multiplexercomprises at least one of a Space Division Multiple Access (SDMA)combiner or equalizer, a maximal ratio combining (MRC) combiner orequalizer, a minimum mean squared error (MMSE) combiner or equalizer, anEigen Beam Forming (EBF) combiner or equalizer, a receive beam forming(BF) combiner or equalizer, a Zero Forcing (ZF) combiner or equalizer, achannel estimator, a Maximal Likelihood (DL) detector, an InterferenceCanceller (IC), a VBLAST combiner or equalizer, a Discrete FourierTransformer (DFT), a Fast Fourier Transformer (FFT), or an Inverse FastFourier Transformer (IFFT).
 13. The backhaul radio of claim 1, whereinthe frequency selective receive path channel multiplexer comprises: aplurality of cyclic prefix removers, wherein each cyclic prefix removeris configured to discard a fraction of an overall number of sampleswithin one or more blocks of a plurality of blocks of samples from arespective one of the plurality of receive chain output signals toproduce a respective cyclic prefix removed one or more blocks ofsamples, said fraction corresponding to a known cyclic prefix length fora plurality of second transmit symbol streams expected to be comprisedwithin the plurality of receive chain output signals; a plurality ofrespective complex Discrete Fourier Transformers coupled to eachrespective cyclic prefix remover, wherein each complex Discrete FourierTransformer is configured to decompose the respective cyclic prefixremoved one or more blocks of samples into a respective set of receivechain frequency domain subchannel samples; and a plurality of receivechannel equalizers coupled to the plurality of respective complexDiscrete Fourier Transformers, wherein each receive channel equalizer isconfigured to produce a set of channel-equalized frequency domainestimates representative of a respective one of the plurality of secondtransmit symbol streams by applying respective stream-specific andchain-specific receive weights to the respective sets of receive chainfrequency domain subchannel samples; wherein said respectivestream-specific and chain-specific receive weights applied to therespective sets of receive chain frequency domain subchannel samplesvary with relative frequency domain subchannel position within suchsets.
 14. The backhaul radio of claim 13, further comprising: a channelequalizer coefficients generator, wherein the channel equalizercoefficients generator is configured to determine the respectivestream-specific and chain-specific receive weights based at least uponcomparison of certain sets of receive chain frequency domain subchannelsamples with certain expected blocks of known frequency domainsubchannel samples expected to be present at certain times within theplurality of receive chain output signals.
 15. The backhaul radio ofclaim 13, further comprising: a plurality of complex Inverse DiscreteFourier Transformers, wherein each complex Inverse Discrete FourierTransformer is configured to compose a respective one of the pluralityof receive symbol streams from respective sets of channel-equalizedfrequency domain estimates representative of the respective one of theplurality of second transmit symbol streams.
 16. The backhaul radio ofclaim 15, wherein each of the plurality of complex Inverse DiscreteFourier Transformers is implemented by a structure executing a complexInverse Fast Fourier Transform (IFFT), and wherein each of the pluralityof complex Discrete Fourier Transformers is implemented by a structureexecuting a complex Fast Fourier Transform (FFT).
 17. The backhaul radioof claim 13, wherein each of the plurality of receive channel equalizerscomprises a number of complex multipliers corresponding to a number ofthe plurality of receive chain output signals, and a combiner.
 18. Thebackhaul radio of claim 1, wherein each of the plurality of transmitsymbol streams comprises at least a plurality of blocks of symbols andwherein the transmit path channel multiplexer is a non-frequencyselective transmit path channel multiplexer comprising: a plurality ofcyclic prefix adders, wherein each cyclic prefix adder is configured toadd a fraction of an overall number of samples within one or more blocksof a plurality of blocks of samples corresponding to a respective one ofthe plurality of transmit chain input signals, said fractioncorresponding to a pre-determined cyclic prefix length; and a pluralityof transmit channel equalizers, wherein each transmit channel equalizeris configured to produce one or more blocks of non-frequency selective,channel-equalized samples corresponding to a respective one of theplurality of transmit chain input signals by applying respective sets ofthe stream-specific and chain-specific transmit beamforming weights tocorresponding blocks of symbols from the plurality of transmit symbolstreams; wherein a particular one of said stream-specific andchain-specific transmit beamforming weights is invariant with respect toa relative symbol position within said blocks of symbols; wherein afirst number of the plurality of transmit chain input signals exceeds asecond number of the plurality of transmit symbol streams; and wherein anumber of the plurality of cyclic prefix adders and of the plurality oftransmit channel equalizers corresponds to the first number.
 19. Thebackhaul radio of claim 18, wherein each respective one of the pluralityof transmit channel equalizers is coupled to an input of a respectiveone of the plurality of cyclic prefix adders.
 20. The backhaul radio ofclaim 18, wherein each of the plurality of transmit channel equalizerscomprises a number of complex multipliers corresponding to the secondnumber, and a combiner.
 21. The backhaul radio of claim 18, wherein thestream-specific and chain-specific transmit beamforming weights aredetermined at a receiver comprised within at least one of the backhaulradio or the one or more other backhaul radios.
 22. The backhaul radioof claim 21, wherein the receiver that determines the stream-specificand chain-specific transmit beamforming weights further comprises: achannel equalizer coefficients generator, wherein the channel equalizercoefficients generator is configured to determine the respectivestream-specific and chain-specific transmit beamforming weights based atleast upon comparison of certain signals at the receiver with certainexpected signals expected to be present at certain times.
 23. Thebackhaul radio of claim 18 wherein the stream-specific andchain-specific transmit beamforming weights are determined in order toimprove either a signal to interference and noise ratio (SINR) or asignal to noise ratio (SNR).
 24. The backhaul radio of claim 20 whereineach of the stream-specific and chain-specific transmit beamformingweights comprises at least a real branch component and an imaginarybranch component.
 25. The backhaul radio of claim 18 wherein each of thestream-specific and chain-specific transmit beamforming weightscomprises at least one of an amplitude component or a phase component.26. The backhaul radio of claim 1, wherein the exchanging of one or moredata interface streams with one or more other backhaul radios is via atleast a Single-Carrier Frequency Domain Equalization (SC-FDE) modulationformat.
 27. The backhaul radio of claim 1, wherein the exchanging of oneor more data interface streams with one or more other backhaul radios isvia at least an Orthogonal Frequency Division Multiplexing (OFDM)modulation format.
 28. The backhaul radio of claim 1, wherein each ofthe plurality of transmit symbol streams comprises at least a pluralityof blocks of symbols and wherein the transmit path channel multiplexeris a non-frequency selective transmit path channel multiplexercomprising: a plurality of cyclic prefix adders, wherein each cyclicprefix adder is configured to add a fraction of an overall number ofsamples within one or more of a plurality of blocks of samplescorresponding to a respective one of the plurality of transmit chaininput signals, said fraction corresponding to a pre-determined cyclicprefix length; wherein the non-frequency selective transmit path channelmultiplexer maps the plurality of blocks of symbols for each respectiveone of the plurality of transmit symbol streams to the plurality ofblocks of samples corresponding to each respective one of the pluralityof transmit chain input signals.
 29. The backhaul radio of claim 1,wherein a total number of directional antenna sub-arrays comprisedwithin the plurality of directional antenna arrays exceeds either orboth of a number of the plurality of receive symbol streams or a numberof the plurality of transmit symbol streams.
 30. The backhaul radio ofclaim 1, wherein a total number of directional antenna sub-arrayscomprised within the plurality of directional antenna arrays exceedseither or both of a number of the plurality of receive chain outputsignals or a number of the plurality of transmit chain input signals.31. The backhaul radio of claim 1, wherein the plurality of directionalantenna arrays are arranged on a plurality of facets with one or moredirectional antenna arrays per facet, and wherein each facet ishorizontally offset relative to at least one other facet.
 32. Thebackhaul radio of claim 1, wherein the plurality of directional antennaarrays are arranged on a plurality of facets with one or moredirectional antenna arrays per facet, and wherein each facet isvertically offset relative to at least one other facet.
 33. The backhaulradio of claim 1, wherein an antenna pattern beamwidth of each of thedirectional antenna sub-arrays is narrower in elevation than an antennapattern beamwidth of each of the directional antenna sub-arrays inazimuth.
 34. The backhaul radio of claim 1, wherein each of theplurality of directional antenna arrays comprises two directionalantenna sub-arrays.
 35. The backhaul radio of claim 34, wherein a firstone of the two directional antenna sub-arrays in each of the pluralityof directional antenna arrays has a first polarization, and wherein asecond one of the two directional antenna sub-arrays in each of theplurality of directional antenna arrays has a second polarization thatis substantially orthogonal to the first polarization.
 36. The backhaulradio of claim 35, wherein each of the plurality of directional antennaelements is one of a patch antenna element or a printed dipole antennaelement.
 37. The backhaul radio of claim 36, wherein said first one ofthe two directional antenna sub-arrays in each of the plurality ofdirectional antenna arrays comprises a plurality of patch antennaelements with each said patch antenna element excited in the firstpolarization, and wherein said second one of the two directional antennasub-arrays in each of the plurality of directional antenna arrayscomprises the same said plurality of patch antenna elements with eachsaid patch antenna element excited in the second polarization.
 38. Thebackhaul radio of claim 37, wherein each of said plurality of patchantenna elements further comprises a first orthogonal probe feed forexcitation in the first polarization for said first one of the twodirectional antenna sub-arrays in each of the plurality of directionalantenna arrays, a second orthogonal probe feed for excitation in thesecond polarization for said second one of the two directional antennasub-arrays in each of the plurality of directional antenna arrays, and ashorting pin at the center of each said patch antenna element.
 39. Thebackhaul radio of claim 1, further comprising: one or moreprobe-in-space antenna elements, wherein at least one of the one or moreprobe-in-space antenna elements has a wider antenna pattern beamwidth inazimuth than an antenna pattern beamwidth in azimuth for any of thedirectional antenna sub-arrays; wherein at least one of the one or moreprobe-in-space antenna elements can be utilized for Dynamic FrequencySelection (DFS) that includes detection of radar systems.
 40. Thebackhaul radio of claim 1, wherein each of the plurality of directionalantenna sub-arrays is comprised of one or more columns of a plurality ofpatch antenna elements.
 41. The backhaul radio of claim 40, wherein anumber of columns for the one or more columns of a plurality of patchantenna elements is less than a number of patch elements within each ofthe one or more columns of a plurality of patch antenna elements. 42.The backhaul radio of claim 1, wherein each of the directional antennasub-arrays comprised within each of the plurality of directional antennaarrays is oriented with respect to an azimuthal antenna pattern at asame azimuthal direction relative to at least one other of thedirectional antenna sub-arrays within its respective one of theplurality of directional antenna arrays and is oriented with respect toan azimuthal antenna pattern at a different azimuthal direction relativeto at least one directional antenna sub-array within at least one otherof the plurality of directional antenna arrays.